I. INTRODUCTION
DC–DC pulse-width modulation (PWM) power converters are widely used in electrical and electronic systems, including renewable energy systems. Switching frequency is often increased to reduce the size and weight of power converters while increasing the power density. However, higher switching frequency leads to more switching losses and electromagnetic emissions. Soft switching techniques are indispensable to overcoming these issues [1]–[7].
Zero-current-transition (ZCT) methods are desirable approach in high power applications [8]–[14], particularly when IGBTs are used as the main switch. As the current becomes zero before the turning off of the switch, the IGBT current tailing losses are alleviated [15]. The ZCT converter proposed in [8] exerts no additional current stress on switches, but the turn-on of the main and auxiliary switches is hard switching, which decreases the efficiency. In [9], ZCS condition is provided for the switches by means of a high frequency resonant network; however, the capacitive losses of the MOSFET output capacitor at turn-on led to additional losses. An improved ZCT converter was introduced in [10] with reduced conduction losses, but the voltage stress of main diode is twice the output voltage. A comparative study on ZCT converters was presented in [11], and an improved ZCT boost converter was introduced; this converter transfers circulating energy to the output via a transformer, thereby increasing efficiency. However, this improved converter comprises more components than other ZCT converters. In [16], three coupled inductors were employed in a ZCS power factor correction boost converter to limit the reverse-recovery current of the output diode. However, five extra diodes are used in the snubber circuit, which increase the cost and the weight.
In [4], coupled inductors were employed to achieve a high voltage gain; however, the main and auxiliary switches operate under hard switching conditions, leading to higher switching losses and lower efficiency. A step-up converter proposed in [17] utilizes a snubber circuit comprising coupled inductors; this structure results in a ripple-free output current, but both switches are turned off under hard switching conditions.
In addition to efficiency, soft-switching conditions, and switch stresses, the electromagnetic interference (EMI) caused by power converters should be considered [18]–[21]. Several ZCT converters, such as those proposed in [12] and [13], exhibit high di/dt due to the reverse recovery of the main diode, which could generate more conducted electromagnetic emissions.
In the present work, a new family of ZCT PWM converters that use coupled inductors is introduced. In the proposed converters, the main and auxiliary switches turn on with ZCT and turn off with both ZCT and ZVT. Moreover, all diodes operate with soft switching condition, and the voltage stress of the main diode is less than twice the output voltage.
Providing ZCS condition at turn-off instant for both switches alleviates the tailing current loss of the IGBT. Furthermore, the lower conduction loss and cost of the IGBT with respect to MOSFET at high voltages makes the proposed auxiliary circuit suitable for high power applications. As the proposed converters have both ZCS and ZVS features, switching losses are almost zero, and they can operate at a higher switching frequency in comparison with their hard switching counterparts.
Utilizing coupled inductors in the proposed converters offers some advantages, such as providing ZVS condition for the main switch and diodes and lowering the current stresses of the switches. In the proposed family, the circulating energy of the auxiliary circuit is transferred to the output by the coupled inductors, thereby increasing overall efficiency. Note that the leakage inductance of the coupled inductors is absorbed by the resonant network and provides ZCS condition for the main switch at the turn-off instant.
The proposed ZCT converters have almost continuous and non-pulsating current waveforms. Moreover, the ZCT condition at turn-on and ZCT–ZVT at turn-off instant result in lower di/dt and dv/dt. Accordingly, the proposed family could potentially improve electromagnetic compatibility (EMC) [20]. Therefore, in addition to the efficiency and soft-switching issues, conducted EMI is evaluated in this work.
This paper is organized as follows. From the proposed converter family, the analysis of the boost topology and its theoretical waveforms are presented in Section II. The design procedure of the proposed converter is described in Section III. The experimental results of the converter prototype that confirm the theoretical analysis are presented in Section IV. The conducted electromagnetic emissions of the proposed converter are measured and compared with those of a similar ZCT boost converter in Section V. The topology variation of the proposed converter is detailed in Section VI, and the concluding remarks are given in Section VII.
II. CIRCUIT DESCRIPTION AND OPERATION
The circuit configuration of the proposed ZCT boost converter is shown in Fig. 1. The circuit components include the main inductor Lin, coupled inductors L1 and L2, output capacitor CO, resonance capacitor Cr, main switch S1, auxiliary switch Sa and its body diode D1, and two other diodes, DO and D2. The total leakage inductance of the coupled inductors is shown by Llk. To simplify the analysis, Lin and CO are assumed to be sufficiently large to be modeled as a constant current and voltage source Iin and VO, respectively. The turn ratio of the coupled inductors is N2 /N1=n, i.e., L2=n2L1. The proposed converter has eight operation intervals, as shown in Fig. 2.
Fig. 1.Proposed ZCT PWM boost converter.
Fig. 2.Equivalent circuit for each operation interval of the proposed boost converter. (a) [t0 – t1], (b) [t1 – t2], (c) [t2 – t3], (d) [t3 – t4], (e) [t4 – t5], (f) [t5 – t6], (g) [t6 – t7], (h) [t7 – t8].
The theoretical waveforms are illustrated in Fig. 3. Before the first interval, all switches are assumed to be turned off, and Iin flows through the main diode DO to the output; thus, Cr voltage is equal to VO.
Fig. 3.Key waveforms of the proposed converter.
Interval 1: [t0−t1] (Fig. 2a): At the beginning of this mode, S1 is turned on under ZCS condition because of the presence of L1. According to the following equation, the current of S1 increases linearly to Iin, which prepares the condition for DO to turn off under ZCS.
At the end of this interval, IS1 reaches Iin, and DO is turned off under ZCS.
Interval 2: [t1–t2] (Fig. 2b): In this mode, Iin flows through S1. This interval is identical to any PWM boost converter when the switch is on.
Interval 3: [t2–t3] (Fig. 2c): At t2, Sa is turned on under ZCS, and a resonance among L1, Llk, and Cr begins through S1 and Sa. The current and voltage equations are as follows:
where
At the end of this interval, IS1 reaches I1, and Cr discharges to −VO/n.
Interval 4: [t3–t4] (Fig. 2d): When VCr reaches −VO/n, D2 starts to conduct and as a result of the coupling between L1 and L2, L1 voltage is maintained at −VO/n. The resonance between Llk and Cr continues through S1 and Sa. The voltage and current equations for this interval are given below.
where
At the end of this interval, IS1 decreases to Iin, Cr discharges to VC1, and ISa becomes zero.
Interval 5: [t4–t5] (Fig. 2e): The resonance between Llk and Cr continues through S1 and the body diode of Sa. Hence, Sa can be turned off under ZCS. In this interval, S1 and D2 currents and Cr voltage can be calculated with (6), (7), and (8), respectively. At the end of this interval, IS1 becomes zero.
Interval 6: [t5–t6] (Fig. 2f): At t5, S1 current becomes zero and can be turned off under ZCT. As a result of the existence of Cr and based on (14), S1 voltage rises linearly, which is considered as ZVT. Consequently, S1 turning off is ZVZCT. The descriptive equations are written as follows:
At the end of this interval, Cr voltage reaches VO. The duration of this mode and the maximum voltage stress of S1 are
Interval 7: [t6–t7] (Fig. 2g): At the beginning of this interval, Cr voltage reaches VO; hence, the body diode of Sa is turned off and DO can be turned on under ZVS, so Iin flows through DO. When the body diode of Sa is turned off, the voltage of Sa remains zero, and there is a delay between the turn-off instant of the body diode and the voltage rise-up of the auxiliary switch. As a result, Sa is turned off under the ZVZCT condition. In this interval, L2 current decreases linearly to zero, and D2 is turned off under ZCS.
Interval 8: [t7–t8] (Fig. 2h): This operating interval is equal to the turn-off state of regular boost converters, in which Iin flows through DO to the output.
III. DESIGN PROCEDURE
A. Design of Component Values
The input inductor and output capacitor of the converter can be designed as regular PWM converters. L1 is a snubber inductor that provides ZCS condition for the main switch at turn-on instant. So it can be designed according to [22].
where Vsw is the switch voltage before turn-on instant, Isw is the switch current after turn-on, and tr is the switch current rise time. Cr provides ZCS condition for main switch at turn-off instant. According to (6), the following equation should be satisfied:
According to (15), increasing Cr also increases the duration of the sixth interval, the transition time of the main switch at turn-off instant, and, consequently, the conduction losses.
To determine n, the maximum allowable duty cycle should be considered. In this converter, the maximum duty cycle is limited by the discharge time of L2 in the sixth and seventh intervals. Thus, by using (12), the maximum allowable duty cycle can be calculated as follows:
According to (20), a large n results in a decreasing maximum allowable duty cycle and more voltage stress across D2. However, (16) indicates that larger n leads to lower voltage stress across the main switches. On the basis of (20) and the aforementioned issues, the proper turn ratio (n) can be selected.
B. Power and Frequency Ranges
The proposed converter can be used in various power ranges, as well as other regular ZCT PWM converters and their hard-switching counterparts. In high voltage applications, the voltage stress of D2, which is equal to (n+1) VO, is increased considerably. However, this limitation is overcome by decreasing n or using series diodes rather than a costly high voltage diode.
In soft-switching PWM converters, the resonant periods must be negligible in comparison with the switching period. The maximum switching frequency can be selected ten times smaller than f1, but in practice, the switch speed restricts the switching frequency. As a result of the soft-switching condition provided for all semiconductor devices, switching losses become almost zero. Consequently, if the selected resonant frequency is sufficiently high, maximum switching frequency is limited by the operation frequency of the switches, which can be reach a few hundred kHz.
C. Auxiliary Switch Timing
The timing relations between the gate pulse of the main switch and the auxiliary switch are presented in this section. Fig. 4 shows the current and gate pulse waveforms of switches in turn-off instant. According to Fig. 4, before S1 is turned off, the auxiliary switch is turned on, leading to a resonance between L1, Llk, and Cr; this resonance provides ZCS condition for the main switch at turn-off instant. The gate pulse of S1 should be removed when its current becomes zero.
Fig. 4.Waveforms of the current and gate pulse of switches at turn OFF instant.
The duration in which the auxiliary switch conducts is shown as α in Fig. 4. According to Figs. 3 and 4 and using (2) and (6), α can be calculated as
β is the time length for decreasing the S1 current from Iin to zero, as shown in Fig. 4. Using (6), β is computed as follows:
During λ, Cr is charged linearly by Iin until its voltage reaches VO. The duration of λ is Δt6 according to (15).
To determine the duty cycle of Sa, the maximum and minimum allowable ton of VGS2 should be considered. According to Fig. 4, by elapsing α, Sa current reaches zero, then its body diode begins conducting. Therefore, the minimum ton is α.
The body diode of Sa continues to conduct during β and λ; hence, the gate pulse of Sa should be removed during this time. The shaded area in Fig. 4 shows the appropriate time for removing the gate pulse of Sa. Accordingly, on-time of Sa can be calculated as
According to Fig. 4, when α + β elapses, the current of S1 becomes zero and it is the proper time for turning off S1.
IV. EXPERIMENTAL RESULTS
A prototype of the proposed ZCT boost converter is implemented with 50 V input and 100 V output. Table I shows the key parameters of the experimental prototype. A photograph of the implemented prototype is shown in Fig. 5. The experimental waveforms of the main switch, auxiliary switch, and diodes D2 and DO shown in Fig. 6–8 confirm the theoretical analysis. Fig. 6 shows that the main switch turns on under ZCS and turns off under simultaneous ZCS and ZVS. To clarify the ZCS condition at turn-on, Fig. 6(b) shows the waveforms of the main switch at turn-on instant.
TABLE IKEY PARAMETERS OF EXPERIMENTAL PROTOTYPE
Fig. 5.Photograph of the implemented prototype.
Fig. 6.Measured voltage (top) and current (bottom) of the main switch S1. (a) At one switching cycle (vertical axis: 50 V/div or 5 A/div, horizontal axis:1 μs/div). (b) At turn-on instant (vertical axis: 25 V/div or 1.5 A/div, horizontal axis: 250 ns/div).
In the ZCT converters of [23] and [24], the coupled inductors provide soft switching, but the main switch only turns off under ZVS. Consequently, the current of the leakage inductor creates voltage and current spikes at the turn-off instant. In the proposed converter, the main switch turns off under ZCS and ZVS; thus, the current of the leakage inductor discharges completely before turn-off, and perfect ZCZVS is achieved.
Fig. 7(a) illustrates the voltage and current waveforms of the auxiliary switch. Fig. 7(b) indicates that the auxiliary switch turns on under ZCS and turns off under ZCZVS. Fig. 8(a) shows that D2 turns on with ZVS and ZCS and turns off with ZCS. The voltage stress of D2 can be lowered by decreasing n, but doing so increases the switch voltage stress.
Fig. 7.Measured voltage (top) and current (bottom) of the auxiliary switch Sa. (a) At one switching cycle (vertical axis: 50 V/div or 3 A/div, horizontal axis:1 μs/div). (b) At turn-on instant (vertical axis: 50 V/div or 2 A/div, horizontal axis: 250 ns/div).
Fig. 8(b) shows the voltage and current waveforms of the main diode DO. There is no current stress on DO, and its voltage stress is less than that of the main diodes in [10], [15] and is less than 2 VO.
Fig. 8.Measured voltage (top) and current (bottom) of (a) diode D2 (vertical axis: 150 V/div or 5 A/div, horizontal axis: 1 μs/div) and (b) main diode (vertical axis: 50 V/div or 5A/div, horizontal axis: 1 μs/div).
The efficiency curve of the proposed converter and that of ZCT boost converter of [25] are compared using the PSpice software (Fig. 9). The efficiency of the proposed converter is 2.5% greater than that of the ZCT boost of [25] and reaches 97% under full load. The proposed converter achieves a higher efficiency because of the better switching conditions and transfer of circulating energy to the output.
Fig. 9.Efficiency of ZCT boost converters (proposed and [25]).
The proposed converter and three other ZCT PWM boost converters are compared in Table II. There are fewer extra elements in [8], but its main and auxiliary switches turn on under hard switching, thus decreasing efficiency as power and frequency increase. In [11], circulating energy was reduced and efficiency was improved, but the number of components used was relatively large.
TABLE IICOMPARISON OF ZCT PWM CONVERTERS
The proposed converter provided better switching conditions than that in [25] for switches and diodes. The current stress of switches and circulating energy decrease considerably because of the coupling effect, which leads to lower losses and higher efficiency. However, the number of the components for the proposed converter is equal to those in [25]; hence, all advantages are achieved without increasing weight or cost. Table III presents the semiconductor losses of the proposed converter with the key parameters mentioned in Table I according to the formula of each component loss [22]. The losses are estimated using component datasheets, whereas the average and RMS values are obtained from the simulation results.
TABLE IIISEMICONDUCTOR LOSSES IN PROPOSED ZCT BOOST CONVERTER
V. CONDUCTED EMI MEASUREMENT
This section presents the experimental results of the conducted EMI measurement for the proposed converters and ZCT boost [25]. A line impedance stabilization network (LISN) according to CISPR22 standard is used to measure conducted electromagnetic emissions [26]. The conducted EMI spectrums are measured using a GW-INSTEK spectrum analyzer (peak detection mode) (as shown in Fig. 10 and Fig. 11 for 150 kHz- 30 MHz.
Fig. 10.Conducted EMI measurement of proposed ZCT boost converter (vertical axis: 20–100 dBμV, horizontal axis: 0.15–30 MHz).
Fig. 11.Conducted EMI measurement of ZCT boost converter in [25] (vertical axis: 20–100 dBμV, horizontal axis: 0.15–30 MHz).
Figs. 10 and 11 show that the main EMI peaks of the proposed and ZCT converters [25] are approximately 82 dBμV at 7 MHz and 90 dBμV at 9.7 MHz, respectively. Consequently, the main EMI peak of the proposed converter is 8 dBμV lower than the main EMI peak of [25] because of the soft switching conditions provided for all semiconductor devices. The conducted electromagnetic emissions of the proposed and ZCT boost converters [25] are compared in Fig. 12 under various frequency ranges. The effect of the proposed ZCT–ZVT method on EMI reduction is significant at frequencies up to 21 MHz. Although the proposed and mentioned converters provide ZCS that leads to lower di/dt, the proposed converter provides ZVS, which also reduces dv/dt. Thus, the proposed boost converter achieves better performance in terms of EMC because of the lower main EMI peak.
Fig. 12.Comparison of conducted EMI peaks in various frequency ranges (experimental results).
VI. OTHER ZCT PWM CONVERTERS
The proposed ZCT method can be applied to other basic PWM converters, such as buck, buck-boost, Cuk, SEPIC, and zeta converters (Fig. 13). In these topologies, the operation intervals are similar to those of the boost converter in Section II.
Fig. 13.Proposed family of ZCT PWM converters: (a) buck, (b) buck-boost, (c) Cuk, (d) SEPIC, and (e) zeta.
VII. CONCLUSION
A new family of ZCT PWM converters is introduced. All semiconductor devices in the proposed converters operate under soft-switching conditions. On the basis of this family, the boost topology is selected. The analysis of the interval modes shows that the main switch turns on with ZCS and turns off with ZCT and ZVT. Moreover, the auxiliary switch turns on under ZCT and turns off under ZVZCT. The additional benefits of the proposed converters included low circulating energy and current stress. A ZCT boost converter from the proposed family is designed and implemented with the proposed design procedure. The experimental results of the prototype confirm the theoretical analysis. That is, the proposed converter achieves a higher efficiency and lower conducted emissions in comparison with another ZCT boost converter.
References
- C. M. Wang, C. H. Lin, and C. H. Cheng, "Analysis, design and realization of a zero-current transition pulse-width modulation interleaved boost power factor correction converter with a zero-current transition auxiliary circuit," IET Power Electronics, Vol. 8, No. 9, pp. 1777-1785, Sep. 2015. https://doi.org/10.1049/iet-pel.2014.0938
- Q. Lou, Y. Zhang, P. Sun, and L. Zhou, "An active clamp high step-up boost converter with a coupled inductor," Journal of Power Electronics, Vol. 15, No. 1, pp. 86- 95, Jan. 2015. https://doi.org/10.6113/JPE.2015.15.1.86
- X. Wei, C. Lou, H. Nan, and Y. Wang, "A simple structure of zero-voltage switching (ZVS) and zero-current switching (ZCS) buck converter with coupled inductor," Journal of Power Electronics, Vol. 15, No. 6, pp. 1480-1488, Nov. 2015. https://doi.org/10.6113/JPE.2015.15.6.1480
- L. S. Yang and C. C. Lin, "Analysis and implementation of a DC-DC converter for hybrid power supplies systems," Journal of Power Electronics, Vol. 15, No. 6, pp. 1438-1445, Nov. 2015. https://doi.org/10.6113/JPE.2015.15.6.1438
- H. F. Xiao, K. Lan, B. Zhou, L. Zhang, and Z. Wu, "A family of zero-current-transition transformerless photovoltaic grid-connected inverter," IEEE Trans. Power Electron., Vol. 30, No. 6, pp. 3156-3165, Jun. 2015. https://doi.org/10.1109/TPEL.2014.2337513
- J. H. Kim, D. Y. Jung, S. H. Park, C. Y. Won, Y. C. Jung, and S. W. Lee, "High efficiency soft-switching boost converter using a single switch," Journal of Power Electronics, Vol. 9, No. 6, pp. 929-939, Nov. 2009.
- K. S. Bin Muhammad and D. D.-C. Lu, "ZCS bridgeless boost PFC rectifier using only two active switches," IEEE Trans. Ind. Electron., Vol. 62, No. 5, pp. 2795-2806, May 2015. https://doi.org/10.1109/TIE.2014.2364983
- C. E. Kim, E. S. Park, and G. W. Moon, "New zero-current-transition (ZCT) circuit cell without additional current stress," Journal of Power Electronics, Vol. 3, No. 4, pp. 215- 223, Oct. 2003.
- M. Jabbari, S. Sharifi, and G. Shahgholian, "Resonant CLL non-inverting buck-boost converter," Journal of Power Electronics, Vol. 13, No. 1, pp. 1-8, Jan. 2013. https://doi.org/10.6113/JPE.2013.13.1.1
- H. S. Park and B. H. Cho, "Improved zero-current-switching (ZCS) PWM switch cell with minimum additional conduction losses," Journal of Power Electronics, Vol. 1, No. 2, pp. 71-77, Mar. 2001.
- P. Das and G. Moschopoulos, "A comparative study of zero-current-transition PWM converters," IEEE Trans. Ind. Electron., Vol. 54, No. 3, pp. 1319-1328, Jun. 2007. https://doi.org/10.1109/TIE.2007.891663
- H. Mao, F. C. Y. Lee, X. Zhou, H. Dai, M. Cosan, and D. Boroyevich, "Improved zero-current-transition converters for high-power applications," IEEE Trans. Ind. Appl., Vol. 33, No. 5, pp. 1220-1232, Sep./Oct. 1997. https://doi.org/10.1109/28.633800
- E. Adib and H. Farzanehfard, "Family of isolated zero current transition PWM converters," Journal of Power Electronics, Vol. 9, No. 2, pp. 156-163, Mar. 2009.
- M. Jabbari, H. Farzanehfard, and G. Shahgholian, "Isolated topologies of switched-resonator converters," Journal of Power Electronics, Vol. 10, No. 2, pp. 125- 131, Mar. 2010. https://doi.org/10.6113/JPE.2010.10.2.125
- E. Adib and H. Farzanehfard, "Family of zero current transition PWM converters," IEEE Trans. Ind. Electron., Vol. 55, No. 8, pp. 3055- 3063, Aug. 2008. https://doi.org/10.1109/TIE.2008.922597
- H. S. Kim, J. W. Baek, M. H. Ryu, J. H. Kim, and J. H. Jung, "Passive lossless snubbers using the coupled inductor method for the soft switching capability of boost PFC rectifiers," Journal of Power Electronics, Vol. 15, No. 2, pp. 366- 377, Mar. 2015. https://doi.org/10.6113/JPE.2015.15.2.366
- S. J. Kim and H. L. Do, "Soft-switching step-up converter with ripple-free output current," IEEE Trans. Power. Electron., Vol. 31, No. 8, pp. 5618- 5624, Aug. 2016. https://doi.org/10.1109/TPEL.2015.2493148
- M. Mohammadi, E. Adib, and M. R. Yazdani, "Family of soft-switching single-switch PWM converters with lossless passive snubber," IEEE Trans. Ind. Electron., Vol. 62, No. 6, pp. 3473- 3481, Jun. 2015. https://doi.org/10.1109/TIE.2014.2371436
- M. R. Yazdani, S. Rahmani, and M. Mohammadi, "A ZCT double-ended flyback converter with low EMI," Journal of Power Electronics, Vol. 15, No. 3, pp. 602- 609, May 2015. https://doi.org/10.6113/JPE.2015.15.3.602
- M. R. Yazdani, H. Farzanehfard, and J. Faiz, "Classification and comparison of EMI mitigation techniques in switching power converters - a review," Journal of Power Electronics, Vol. 11, No. 5, pp.767-777, Sep. 2011. https://doi.org/10.6113/JPE.2011.11.5.767
- M. R. Yazdani, H. Farzanehfard, and J. Faiz, "EMI analysis and evaluation of an improved ZCT flyback converter," IEEE Trans. Power Electron., Vol. 26, No. 8, pp. 2326- 2334, Aug. 2011. https://doi.org/10.1109/TPEL.2010.2095884
- N. Mohan, T. M. Undeland, and W. P. Robbins, Power Electronics, Wiley, 3rd edition, 2003.
- M. R. Amini and H. Farzanehfard, "Novel family of PWM soft-single-switched dc-dc converters with coupled inductors," IEEE Trans. Ind. Electron., Vol. 56, No. 6, pp. 2108- 2114, Jun. 2009. https://doi.org/10.1109/TIE.2009.2016509
- E. Adib and H. Farzanehfard, "Family of soft-switching PWM converters with current sharing in switches," IEEE Trans. Power Electron., Vol. 24, No. 4, pp. 979- 984, Apr. 2009. https://doi.org/10.1109/TPEL.2008.2008022
- M. Ahmadi, M. R. Mohammadi, E. Adib, and H. Farzanehfard, "Family of non-isolated zero current transition bi-directional converters with one auxiliary switch," IET Power Electronics, Vol. 5, No. 2, pp. 158-165, Feb. 2012. https://doi.org/10.1049/iet-pel.2011.0098
- IEC International Special Committee on Radio Interference - C.I.S.P.R., "Information Technology Equipment-Radio Disturbance Characteristics-Limits and Methods of Measurement," International Electrotechnical Commission, Publication 22, 1997.
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