I. INTRODUCTION
Today, power switching converters are widely used in many applications such as fuel cells [1], LED drivers [2], power factor correction [3] and photovoltaic systems [4]. Among the various switching converters, the flyback topology is widely used in electronic systems due to its benefits such as isolation and a simple structure. The best way to decrease the weight and size of a converter is to increase the switching frequency. Due to the non-ideal behavior of the semiconductor elements used in power converters, increasing the switching frequency may lead to increases in both switching losses and electromagnetic interference (EMI). It may also drive the converter switch out from the safe operating area (SOA). Over the years, many methods have been introduced to diminish these problems. Generally, soft switching techniques including both passive and active methods are commonly used to counteract the repercussions of increasing the switching frequency in PWM converters. Because passive snubber circuits use no additional active components to provide soft switching conditions, they usually do not increase the complexity of the control and gate drive circuits. In [5], a lossless passive snubber cell is introduced which can be engaged on double-ended flyback converters. However, the number of components in each cell is high. In addition, because two snubber cells should be applied for each converter switch, the converter power conversion density is affected. In [6], a lossless passive snubber is suggested for double-ended flyback converters. When the converter switch is on, the current through the snubber inductors freewheels through the converter switches and the two snubber diodes. This causes the conductive losses to increase. In addition, the number of the snubber components is relatively high. In [7], a half bridge interleaved flyback converter is introduced which uses a simple lossless passive snubber cell. Although, the snubber circuit is simple, two snubber cells should be used for the two converter switches. Moreover, two distinct cores must be engaged in this converter. Passive snubbers are usually able to achieve zero voltage and zero current switching conditions at the turning off and on moments, respectively. However, some active methods provide zero voltage and zero current transition (ZVT and ZCT) conditions at the turning on and off instants, respectively [8], [9]. In addition to these methods, forcing the converter to work in critical conduction mode (CRM) can lead to soft switching conditions. Although, working under this condition decreases the switching losses, it increases the conduction losses due to an increased rms current. In addition, the CRM results in a more complicated control circuit than regular converters. Moreover, the switching frequency cannot be constant which results in a non-optimum filter design.
Among the isolated converters, the flyback converter has a low cost and simple structure [10]. Despites, the advantages of the single-switch flyback topology, it has some disadvantages which make it difficult to be used especially in medium power applications. The switch voltage stress and the voltage spikes caused by the transformer leakage inductor (Llk) are two major problems of the conventional single-switch flyback converter. In addition, switching losses and EMI are concerns that should be considered. The double-ended flyback converter clamps switch voltage stress to the input voltage. Furthermore, recycling the Llk energy to the input voltage significantly reduces the voltage spikes across the converter switches. In addition to the switch stress and switching losses reduced by soft switching techniques, the EMI is an important parameter that should be taken into consideration in soft switching converters. Unwanted resonances and other parasitic elements due to an increased component quantity may lead to insufficient EMI reduction in some soft switching converters. In [11], a ZCT single-switch flyback converter with a simple auxiliary cell at the secondary side of the transformer is introduced. The auxiliary switch voltage suffers from voltage spikes at the turn-off instant which affects the electromagnetic emissions [12]. In [13], a double-ended ZCT flyback converter with an auxiliary cell at the secondary side of the transformer is proposed. Although, it is desirable to apply soft switching cells with a minimum need for isolated gate drive circuits, three isolated gate drivers must be used in [13]. In addition, there are considerable high frequency oscillations in the waveforms of the main and auxiliary switches which may result in more common-mode (CM) and differential-mode (DM) EMI. CM EMI is generated with high di/dt and dv/dt across the parasitic elements between the circuit and the chassis (earth). An important CM impedance is the capacitance between the switch drain and the earth (CDE) including the heat-sink capacitance [14]. In the double-ended flyback converter, the drain of the upper switch is connected to the DC input voltage and consequently its CDE is not important from the EMI viewpoint. Due to providing soft switching conditions via the suggested snubber cell, the impact of the lower switch’s CDE on the EMI is limited.
In this paper, a double-ended flyback converter with an active auxiliary cell is introduced which causes the converter switches to be turned off under the ZCT condition. Due to the transformer leakage inductor, the converter switches are turned on under the ZCS condition. In addition, the voltage stress of the main switches is clamped to the input voltage. Clamping the voltage stress leads to lower voltage across the switches. Generally, MOSFETs with a lower voltage have a lower ON resistance. Using these MOSFETs will reduce the conductive losses. In the proposed converter, the EMC can be also improved by providing the ZC condition at the turn off instant. In the proposed converter because the switching frequency is relatively high, the size is reduced. In addition, the switching losses are decreased due to providing soft switching conditions. The auxiliary snubber cell is responsible in obtaining this condition. Moreover, In the proposed topology only one floating gate drive circuit is needed. As a result, in this the case boot strap technique can be used to drive the upper switch to prevent the use of a pulse transformer or an optocoupler.
This paper is arranged into five sections. In section II, the proposed ZCT double-ended flyback converter is discussed in detail. Section III provides an appropriate design procedure for the active auxiliary snubber cell. In order to justify the theoretical discussions about the proposed converter, section IV presents some experimental results. In section V, conducted EMI measurements of the converter are presented. Finally, in section VI the remarkable points of the aforementioned discussions are presented.
II. THE PROPOSED CONVERTER
In Fig. 1, the proposed ZCT double-ended flyback converter is shown. The auxiliary circuit is made up of Lr, Cr, Sa, Da1 and Da2. The proposed converter in each switching period has 10 operating intervals which are discussed in detail below. Fig. 2 shows the equivalent circuits of each interval, and Fig. 3 shows the important theoretical waveforms of the proposed converter. Before explaining the operating modes, it is assumed that the converter switches S1 and S2 along with the auxiliary switch Sa are all off, all of the diodes except DO are off, and VCr is VCr(t0). In this case the transformer magnetizing inductor Lm is transferring its stored energy to the output voltage.
Fig. 1.The proposed ZCT double-ended flyback converter
Fig. 2.The equivalent circuits of each operating mode.
Fig. 3.Theoretical waveforms of the proposed converter.
Interval 1 [t0–t1]: At t0 the converter switches S1 and S2 are simultaneously turned on under the ZC condition, due to the transformer leakage inductor Llk. In this interval, because the current through Llk is lower than ILm, DO remains on and so the voltage of Vin+nVO is placed across Llk which causes its current to increase linearly. The important equations of this interval are as follows:
Interval 2 [t1–t2]: At t1 the current of Llk is equal to the current through Lm. At the beginning of this mode, DO turns off under the ZC condition. The operation of the proposed converter in this interval is the same as that in the conventional flyback converter where Lm is being charged via the input DC voltage. The important equation of this interval is as follows:
Interval 3 [t2–t3]: To turn the main converter switches off under the ZC condition, the auxiliary cell should reduce their currents to zero. Therefore, at t2, Sa is turned on under the ZC condition. Turning Sa on starts a resonance between Lr and Cr. The important equations of this interval are as follows:
where:
Interval 4 [t3–t4]: At t3, after half of the resonance is started in interval 3, the current through Lr wants to flow in the opposite direction. Due to Da2, which is installed in series with Sa, the current cannot flow through Sa. Therefore, Saand Da2 turn off under the ZC condition. Therefore, Da1 turns on under the ZC condition. In this mode the current of Lr and the voltage of Cr can be calculated via equations (4) and (5), respectively.
Interval 5 [t4–t5]: At t4, ILr reaches ILm and causes the current through S1 and S2 to become zero. Then, the current thorough Lr increases more than ILm. When ILr is higher than ILm, the converter switches S1 and S2 can be turned off under the ZCS condition. Turning S1 and S2 off causes the body diodes DS1 and DS2 to turn on under the ZV condition. Thus, the resonance between Cr and Lr continues through the provided pass via DS1 and DS2.
Interval 6 [t5–t6]: At the beginning of this interval, the current through Lr reaches ILm again. As a result, DS1 and DS2 turn off under the ZC condition. In this mode, a new resonance occurs between Cr, Lr, Llk and Lm. Because Lm is very large with respect to Llk and Lr; during this interval, it can be assumed that Cr is charged with a constant current. The important equation of this mode is as follows:
Interval 7 [t6-t7]: At t6, the voltage of Cr reaches -[1+(Llk + Lr)/Lm]nVO. As a result, DO turns on. In this interval a new resonance starts between Llk, Lr and Cr.
where:
Interval 8 [t7-t8]: At t7, the voltage of Cr reaches -(1+Lr/Llk)Vin + (Lr/Llk)nVO . On the other hand, the voltages of the converter switches S1 and S2 become Vin. Thus, DC1 and DC2 turn on under the ZV condition. In this interval, a voltage of Vin-nVO drops across Llk inversely and its current decreases linearly. In addition, in this mode a resonance occurs between Lr and Cr. Under this resonance the current through Lr is reduced. The important equations of this mode are as follows:
Interval 9 [t8-t9]: At t8, ILr reaches zero and Da1 turns off under the ZC condition. In this mode, Llk still has energy. As a result, DC1 and DC2 remain on. During this interval, the current through Llk can be calculated with Equ. (11).
Interval 10 [t9-t10]: At t9, ILlk becomes zero. As a result, DC1 and DC2 turn off under the ZC condition. The operation of the proposed converter in this mode is the same as that in the conventional double-ended flyback converter.
III. DESIGN GUIDELINES
There are two parts to the design of the proposed double-ended flyback converter. The first part is the design of the power section and the second part is the calculation of the values of the snubber components. The first part can be computed in the same way as any conventional flyback converter [15]. In this section a simple approach to choose proper values for the auxiliary cell is introduced. In order to achieve the ZC condition at the turning off instant for the main converter switches, two criteria should be considered. The first criterion is that the resonant period starting in interval 3 must be smaller than the minimum switch on time. Therefore, ω0 should be chosen with equation (14).
where fsw and D are the switching frequency and the minimum duty cycle, respectively.
The second criterion is that the maximum resonant current starting in interval 3 must be larger than ILm at the maximum output power. As a result, equation (15) can be used to calculate Lr.
where is the maximum current of Lm at the maximum output power. In addition, as discussed in section II, at the end of Interval 7, the voltage of Cr is nearly equal to –Vin. Therefore, this quantity is a good estimation of VC(t0). By using Equations (14) and (15), Cr and Lr, can be calculated. In addition, to compute at the maximum output power, the equation below can be used.
where , η and fsw are the nominal power of the converter, the converter efficiency, and the switching frequency. In the worst case, the efficiency while considering an over design can be selected as 0.8.
Another parameter for designing the proposed converter is the selection of appropriate semiconductor elements. The voltage stresses of the main components are shown in Table I. To simplify the calculation of VCr(t0), it is assumed that the resonance between Cr and Lr is complete. Meanwhile, the converter works between Modes 3 to 8. Therefore, VCr(t0) can be obtained by Equation (5). Based on this assumption, its value is around –Vin.
TABLE IVOLTAGE STRESSES OF MAIN SEMICONDUCTOR COMPONENTS
IV. EXPERIMENTAL RESULTS
To assess the theoretical analysis of the proposed converter, a 60W prototype of a ZCT double-ended flyback converter is implemented with a 100 kHz switching frequency. The input and output voltages are 75V and 24V, respectively. The Lm and Llk values are 590μH and 9uH and the turns ratio of the transformer is 2.32. Lr and Cr are chosen as 16μH and 4.7nF based on the design procedure introduced in the previous section. IRF540s are utilized for the switches and a BYV32-200 and UF4001s are used as the output diode DO and snubber diodes, respectively.
Fig. 4 shows the experimental waveforms of the voltage and current of S2 and the voltage of the auxiliary switch Sa. In Fig. 4(a) it is evident that S2 is turned on and off under the ZC condition. Fig. 4(b) shows the voltage and current of the auxiliary switch Sa to demonstrate that the switch acts under the soft switching conditions during transition times. The switch voltage stress of S1 and S2 is clamped to the input voltage which prevents fluctuations after turning the switches off. This reduces the electromagnetic emissions of the converter to [12]. To demonstrate that the soft switching condition for the proposed converter has the ability to improve the converter efficiency, the converter efficiency curve is depicted in Fig. 5. To compare the proposed converter in terms of efficiency, the efficiency curves of the soft switching converter introduced in [6] and the soft switching double-ended flyback converter with are RCD snubber, are also illustrated in Fig. 5. It should be mentioned that the specifications of the mentioned converters are same as the specifications of the proposed converter. It can be seen that the proposed soft switching cell improves the converter efficiency by more than 8% at the nominal output power with respect to the regular converters. It can also be seen that it has better efficiency in low powers than the converter introduced in [6]. In addition, when compared to [13], the proposed ZCT converter has better efficiency. It should be mentioned that the efficiency improvement of the proposed converter can be increased by using a unidirectional switch instead of Sa and Da2 to eliminate Da2 losses.
Fig. 4.(a) Top: S2 voltage, Middle: Sa voltage, Bottom: S2 current. (ver. scale: 2.5 A/div or 50 V/div; time scale: 1 μs/div). (b) Top: Sa voltage, Bottom: Sa current. (ver. scale: 2 A/div or 50 V/div; time scale: 1 μs/div).
Fig. 5.The efficiency curves.
V. CONDUCTED EMIMEASUREMENT
To examine the conducted EMI reduction, the conducted electromagnetic emissions of the proposed converter are measured and compared with those of a conventional double-ended flyback converter. In addition, to show the benefits of the proposed converter from the EMC viewpoint, the soft switching double-ended converter introduced in [13] was made and its electromagnetic emissions were measured. To measure the EMI, a line impedance stabilization network (LISN) based on the CISPR 22 standard is placed between the input of the converter and the input source lines [16]. The conducted EMI spectrum of the proposed ZCT converter, its hard switching counterpart and the ZCT converter in [13] are measured via a HAMEG spectrum analyzer (peak detection mode) as shown in Fig. 6.
Fig. 6.Conducted EMI Measurement of double-ended flyback converters. (a) Hard switching (b) ZCT [13] (c) Proposed ZCT (Vertical axis: 0-100dBμV, Horizontal. axis: 150k-30MHz).
Fig. 6 indicates that the main EMI peak of the proposed converter is about 5 dBμV lower than the main EMI peak of its hard switching counterpart in the total frequency band. In addition, the main EMI peak of the proposed converter is around 7 dBμV lower than the main EMI peak of the ZCT converter introduced in [13]. To compare the conducted EMI in various frequency ranges, the EMI of the proposed converter along with the hard switching counterpart and the suggested converter in [13] are shown in Fig. 7. Comparing these three converters, the conducted EMI levels of the proposed converter are reduced.
Fig 7.Comparison of EMI measurement for various frequency ranges.
To study the effect of output power variations on the conducted EMI levels, the conducted EMI spectrums for various output powers are measured. Although there are small variations in the EMI levels in some frequencies, the main EMI peaks have no considerable changes for various output powers. Since the proposed converter has a lower main EMI peak, it has the potential to meet the EMC standard via simple and cost-effective EMI reduction methods. One of the EMI reduction methods which can be used for this is a Faraday shield. In this technique, the value of the parasitic capacitor between two the sides of the converter transformer is decreased with a piece of aluminum foil placed between the primary and secondary transformer windings [12]. Another simple method to reduce EMI is to use a common-mode choke at the input of the converter. A common-mode (CM) choke provides a high impedance pass at the input of the converter which improves the converter EMI. The EMI spectrum of the proposed converter with a Faraday shield and a CM choke (with two coupled 300 μH inductance) is depicted in Fig. 8.
Fig. 8.EMI reduction after applying Faraday shield and CM choke.
Although the EMI peak of the proposed converter is near the limit of CISPR22 Class A QP (Quasi peak) at low frequencies, this standard is satisfied by the proposed ZCT double-ended flyback converter with two simple EMI reduction techniques. It should be mentioned that the CISPR 22 standard is not satisfied by the regular double-ended converter with these two EMI reduction methods due to its high EMI levels.
VI. CONCLUSION
In this paper, a ZCT double-ended flyback converter is introduced in which all of the semiconductor elements act under soft switching conditions. The performance of the proposed converter is investigated in detail and experimental results are offered to verify the theoretical analysis. The engaged active auxiliary cell improves both the converter efficiency and the EMI. Since the achieved soft switching conditions significantly reduce the converter EMI, the proposed converter can satisfy the CISPR22 standard by simple and cost-effective EMI reduction methods. A Faraday shield and a CM choke (simple EMI reduction methods) are applied to the converter to satisfy the CISPR22 standard. The results of the EMI measurements verify CISPR22 satisfaction after applying these two simple methods.
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