Dual-Output Single-Stage Bridgeless SEPIC with Power Factor Correction

Abstract

This study proposes a dual-output single-stage bridgeless single-ended primary-inductor converter (DOSSBS) that can completely remove the front-end full-bridge alternating current-direct current rectifier to accomplish power factor correction for universal line input. Without the need for bridge diodes, the proposed converter has the advantages of low component count and simple structure, and can thus significantly reduce power loss. DOSSBS has two uncommon output ports to provide different voltage levels to loads, instead of using two separate power factor correctors or multi-stage configurations in a single stage. Therefore, this proposed converter is cost-effective and compact. A magnetically coupled inductor is introduced in DOSSBS to replace two separate inductors to decrease volume and cost. Energy stored in the leakage inductance of the coupled inductor can be completely recycled. In each line cycle, the two active switches in DOSSBS are operated in either high-frequency pulse-width modulation pattern or low-frequency rectifying mode for switching loss reduction. A prototype for dealing with an $85-265V_{rms}$ universal line is designed, analyzed, and built. Practical measurements demonstrate the feasibility and functionality of the proposed converter.

I. INTRODUCTION

The topologies of power factor corrector (PFC) generally contain buck, boost, buck–boost, Zeta, ‘Cuk, single-ended primary-inductor converter (SEPIC), and flyback. Buck PFC can decrease the input voltage to obtain an output voltage less than the peak value of the line voltage [1]. However, zero-crossing distortion degrades power factor. On the contrary, boost PFC can achieve a unity power factor, but its output voltage is higher than its input. Boost PFC cannot deal with the applications of low output voltage, unless it embeds a step-down direct current (dc)/dc stage. Another disadvantage of boost PFC is that power components withstand high-voltage stresses [2], [3]. Buck–boost can obtain an output voltage whose magnitude is either larger or smaller than the input. Nevertheless, the polarity reversal on output and isolated driving requirement will become potential problems [4]. Similar to buck–boost, ‘Cuk and Zeta have the feature of stepping up or down input voltage. However, pulsating input current and high-side driving are required for Zeta, while ‘Cuk still has the polarity reversal problem [5]-[7]. Flyback PFC can resolve the polarity reversal problem and possesses the characteristic of galvanic isolation, but it has the significant drawback of low efficiency [8]-[13]. Compared with the aforementioned step-up/step-down PFCs, the SEPIC type performs better in total harmonic distortion (THD), efficiency, and power factor [14]-[16].

To reduce component count and improve efficiency, a bridgeless structure attracts a great deal of interest in fulfilling power factor correction. Although high efficiency can be achieved in typical bridgeless PFC topologies derived from boost, buck, or buck-boost [17]-[20], the aforementioned drawbacks still exist. Some researchers have proposed a bridgeless PFC with respect to SEPIC configuration [21]-[23]; however, two diodes are needed to accomplish rectification. Given that the two diodes have to block a voltage higher than the mains, the problems of large cut-in voltage and reverse recovery losses remain. In literature [24], the front-end alternating current-dc bridge rectifier is completely done away with, but the step-down property is lost.

To overcome all the mentioned drawbacks, a novel dual-output single-stage bridgeless SEPIC (DOSSBS) with power factor correction is proposed (Fig. 1). Unlike the conventional PFC, the proposed DOSSBS can complete power factor correction without the need for a front-end bridge rectifier, which therefore simplifies converter structure, avoids the problem of power loss on rectifier diode, and decreases component count. DOSSBS is distinguished by the features of single stage, bridgeless, and high efficiency. It can also provide dual individual outputs in a single stage. A 100 W universal line input prototype is built and examined for verification. Experimental results validate the proposed DOSSBS.

Fig. 1.Main power circuit of the proposed DOSSBS.

With respect to practical applications, the proposed DOSSBS can serve as a power supply with power factor correction to drive electric appliances, which need two different levels of source voltage. DOSSBS can power appliances in a single converter, instead of two separate converters, thereby yielding high energy conversion efficiency and low cost. For example, in an intelligent lighting system application, DOSSBS can simultaneously drive light-emitting diodes and provide power for the dimming circuits of the communication interface.

The remainder of this paper is organized as follows. Section II describes the operation principle of the proposed DOSSBS. Section III deals with the design considerations of the converter. Section IV provides practical measurements and a performance comparison with other PFCs. Finally, Section V concludes.

 

II. OPERATION PRINCIPLE

For the operation description of the proposed DOSSBS, the definitions of current direction and voltage polarity are given in Fig. 2. The two active switches alternately operate at a high frequency within an interval of line cycle. During the positive half-line cycle, SW1 is always closed, and SW2 operates at a high frequency. During the negative half-line cycle, SW1 switches in a high frequency, while SW1 is kept in on-state. As the high-frequency switching pattern is not in constant use, the switching loss of the proposed converter can be significantly reduced.

Fig. 2.Representation of voltage polarity and current direction of the proposed converter.

The operation of the converter can be divided into four main modes over one switching period. Figs. 3 and 4 show the corresponding equivalents and conceptual key waveforms when the converter is operated in the positive half cycle respectively. The corresponding mode-equivalents and conceptual key waveforms in the negative cycle are illustrated in Figs. 5 and 6. The converter operation in the positive half cycle is discussed mode by mode below.

Fig. 3.Equivalents during one switching period in the positive half-line cycle. (a) Mode 1, (b) Mode 2, (c) Mode 3, and (d) Mode 4.

Fig. 4.Conceptual waveforms corresponding to the operation modes in the positive half-line cycle.

Fig. 5.Equivalents during one switching period in the negative half-line cycle. (a) Mode 1. (b) Mode 2. (c) Mode 3. (d) Mode 4.

Fig. 6.Conceptual waveforms corresponding to the operation modes in the negative half cycle.

Mode 1 [Fig. 3(a), t0-t1]: Over the entire positive half-line cycle, SW1 is always in on-state, while SW2 is operated in a high switching frequency to control input current. SW2 is turned on at the beginning of mode 1. In mode 1, input voltage is directly connected to the input inductor L1. The current iL1 is linearly built, and the capacitor C2 dumps energy to the primary of the coupled inductor through SW2 and D3. Meanwhile, the capacitor Co2 supplies energy for the load RL2, and the capacitor Co1 for the load RL1. The voltage across the inductor L1 is given by

where vs(t) represents the line voltage, and VDS1,on and VDS2,onstand for the voltage drop on SW1 and SW2 respectively. Supposing that the line voltage is purely sinusoidal and equal to Vmsin(2pflinet), the above equation becomes

where Vm is the amplitude of line voltage, and fline denotes the line frequency. The on-state voltage VDS1,on in Eq. (2) is less than the forward voltage of a rectifier diode. Compared with traditional full-bridge PFCs, the proposed DOSSBS replaces an active switch with two low-frequency rectifier diodes, such that it can significantly decrease conduction loss. The inductor current iL1 can be determined as follows:

Under the boundary mode operation, given that the initial value of the inductor current iL1 (0) is zero, the converter can achieve zero current switching feature at SW2.

In mode 1, the capacitor C2 discharges to the primary magnetizing inductance Lm,pri and the primary leakage inductance Llk,pri of the coupled inductor. The current iD3 can be calculated by

where vC2(t) stands for the voltage across the capacitor C2, VD3,fmeans the forward voltage of the diode D3, and Lpri denotes the measured inductance with respect to the input terminals of the coupled inductor while the secondary is open. Lpri is the sum of Lm,pri and Llk,pri.

Mode 2 [Fig. 3(b), t1-t2]: his mode begins as soon as SW2 is turned off. In mode 2, the energy stored in the inductors L1, Lm,pri, and Llk,pri continues to increase. The voltage across the parasitic capacitor of SW2 also increases, but the capacitors Co1 and Co2still dump energy to the loads RL1 and RL2 respectively. The voltage across SW2 can be expressed as

The voltage vDS2 continuously increases during mode 2. At the moment that vDS2 reaches the magnitude of input voltage, this mode ends, and the polarities of the inductors L1, Lm,pri, and Llk,pri reverse. All the inductors start discharging.

Mode 3 [Fig. 3(c), t2-t3]: During this mode, the inductor L1releases energy to the capacitors C2 and Co2, and the energy stored in the leakage inductance Llk,pri will be recycled to the output through Lm,pri, D2, and D3. Lm,pr dumps energy to Co2 and Co1.

From the equivalent circuit of mode 3, the voltage across the parasitic capacitor of SW2 is clamped at vc2 + Vo2. Thus, the voltage stress of SW2, Vstress, and SW2, can be determined as follows:

The inductor current iL1 can be obtained by

where iL1 (t2) is the initial value of iL1 at t = t2, and VD2,f is the voltage drop on the diode D2. This mode ends when the current of the leakage inductance Llk,pri drops to zero.

Mode 4 [Fig. 3(d), t3-t4]: In mode 4, the inductor L1 continues supplying energy to the capacitors C2 and Co2, while the coupled inductor transmits energy to Co1. The current flowing through Lm,pri linearly decreases, which can be estimated by

In the negative half-line cycle, the roles of SW1 and SW2 exchange. SW1 is operated in high frequency to control input current, while SW2 is kept in on-state over the entire half cycle. The operation of the proposed converter is symmetrical in two half-line cycles of input voltage. The description of operation in the negative half-line cycle is similar to that in the positive which also has four modes. The equivalent circuits and conceptual waveforms are illustrated in Figs. 5 and 6, respectively.

 

III. DESIGN CONSIDERATIONS

A. Equivalent Iron Loss Resistance

1) Vo1 = Vo2 : Minimum switching frequency fsw,min is a key parameter for the design of the input inductor L1. Switching frequency can be estimated by determining the on-time and off-time periods of the active switch. As power factor correction is performed under a constant on-time switching pattern, fsw,min can be obtained after determining the maximum off time. However, a low line input voltage indicates a high on-time interval. Thus, maximum on time Ton,max and maximum off time Toff,max over the range of universal line input should be determined in advance for fsw,min calculation.

Supposing that the range of universal line input is from Vrms,min to Vrms,max, Ton,max then occurs when the line voltage is Vrms,min. Given that output power is equal to the multiplication of input power and converter efficiency, the following equation holds:

where η denotes the converter efficiency, and Po is the output power. Over a half-line cycle, the maximum off time Toff,max appears at the peak of the sinusoidal line voltage. Accordingly, if the two output voltages of the converter are equal, Vo1 = Vo2 = Vo, then

From Eqs. (9) and (10), solving Ton,max and Toff,max obtains

and

The minimum switching frequency is calculated by

Substituting Eqs. (11) and (12) into Eq. (13), we obtain

To determine the value of the input inductor L1, Eq. (14) can be rewritten as

Supposing that η = 0.93, then two output voltages are equal, and the minimum line input voltage is 85 Vrms. Fig. 7 shows the relationship among input inductance, power rating, and output voltage. High power rating requires low input inductance; under a certain power rating, a high output voltage needs large input inductance. Considering that a small input inductance will result in a high switching frequency, the minimum switching frequency should be larger than 20 kHz to avoid audio frequency.

Fig. 7.Relationship between input inductance and output power under different output voltages.

2) Vo1 ≠ Vo2 : The determination of input inductance in Eq. (15) is only suitable for the condition Vo1 = Vo2. If Vo1 ≠ Vo2, then the minimum switch frequencies of SW1 and SW2 will differ. Therefore, two values will be elected as input inductance and are expressed as

and

fsw1,min and fsw2,min denote the minimum switching frequencies of SW1 and SW2 respectively. The smaller one between Lα and Lβ is chosen as the input inductance, that is,

The converter power rating is 100 W, and the minimum line voltage is 85 Vrms. Fig. 8 illustrates the relationship among two output voltages and the minimum switching frequencies of SW1 and SW2. fSW1,min will be larger than fSW2,min when Vo1 > Vo2. On the contrary, fSW1,min is less than fSW2,min when Vo1 < Vo2. If Vo1 = Vo2, both switching frequencies are identical.

Fig. 8.Relationship among the minimum switching frequencies of SW1 and SW2 and dual output voltages.

fSW1,min and fSW2,min vary with input line voltage and output power. For example, η = 0.93, Vo1 = 30 V, Vo2 = 60 V, and input inductance L1 = 670 μH. Fig. 9 shows the relationship among the minimum switching frequencies of the two active switches, input voltage, and output power.

Fig. 9.Relationship between minimum switching frequency and input voltage under different output powers.

B. Design of the Coupled Inductance

The coupling coefficient of a coupled inductor can be evaluated as follows:

Rearranging Eq. (19) yields

When the inductance Lsec is in the output terminals while the primary is open,

where Llk,sec is the leakage inductance of the secondary. The relationship between Llk,pri and Llk,sec is given by

The magnetizing inductance in the secondary Lm,sec also equals the primary magnetizing inductance Lm,pri times the square of turn ratio. The following relationship is then derived:

The terminal voltage of the secondary, VLsec, can be computed by

where VLm,pri is the voltage across the magnetizing inductance of the primary. Given that k is less than unity, the following inequality holds:

Thus, the design for the coupled inductor should meet the following inequality:

C. Coupled Capacitance

The energy-transferred capacitors C1 and C2 are also key components because their values significantly influence input line current. Both capacitors must be in a proper design for their steady-state voltage waveforms to be consistent with the rectified input line voltage, and the low-frequency oscillating with input inductor or coupled inductor can be avoided. In practical consideration, resonant frequency should be larger than line frequency but less than minimum switching frequency, that is,

and

where fr1 and fr2 are the resonant frequencies of L1-C2-Lpri and L1-C1-Lsec respectively. They are calculated as

and

According to Eqs. (29) and (30), the capacitances of C1 and C2 can be calculated by

and

respectively.

D. Output Capacitance

The frequency of output ripple is twice the line frequency. A low output voltage ripple is accompanied by a large output capacitance. Once the output voltage ripple △Vo is specified, the corresponding output capacitance Co can be estimated by

E. Switch Stress

In this converter, the voltage stress across active switch can be estimated by the peak value of the uppermost universal line voltage Vpk,max plus output voltage. Therefore,

and

 

IV. EXPERIMENTAL RESULTS

A prototype is built, simulated, and examined to verify the feasibility of the proposed DOSSBS. In the prototype, the universal line input voltage is over the range of 85-265 Vrms , the line frequency is 60 Hz, the output voltage of ports 1 and 2 are 30 and 60 V respectively, that of port 2 is 60 V, and the converter power rating is 100 W. The key component values are summarized in Table I.

TABLE IKEY COMPONENTS AND VALUES OF THE PROTOTYPE

Fig. 10 shows the measured waveforms of the line voltage vs and the input inductor current iL1 at full load when the line voltage is 110 Vrms. The envelope of iL1 in Fig. 10 is sinusoidal and can be in phase with the line voltage. In the positive half-line cycle, SW2 is operated at a high frequency, but SW1 is always in on-state. Fig. 11 shows the zoomed-in waveforms in the positive half-line cycle. The control signal of SW2 is in a high frequency, and the input inductor current is controlled at a boundary conduction manner. The filtered source current iin is shown in Fig. 12, which illustrates that iin is sinusoidal and in phase with the line voltage. Fig. 13 shows the two output voltages at ports 1 and 2 to demonstrate that both ports can be kept constant at 30 and 60 V under full load. Figs. 14 and 15 present the corresponding waveforms of the step-change transient response of DOSSBS, while the output power at port 1 changes from light to heavy load and from heavy to light load respectively. Both figures indicate that even under step-change loading, DOSSBS still can still feature rapid transient response and sustain stable output voltages. The waveforms of vC1 and vC2 are presented in Fig. 16, which depicts that the positive voltages of vC1 and vC2 are sinusoidal, and the maximum negative voltages equal the output voltages of ports 1 and 2. Fig. 17 presents the voltages across the diodes D1 and D2 when the line voltage vs increases to 90 V. The blocking voltage of D2 is equal to vC2 plus output voltage at port 2, at approximately 150 V. Under the same input voltage of 90 V, the measured waveforms of vD3 and vD4 are shown in Fig. 18. The reversed voltage across D4 is approximately 75 V. The result of harmonic measurement is shown in Fig. 19, which expresses that DOSSBS can meet the standard of IEC 61000-3-2 Class C. The measured THD is 14.8%. Fig. 20 shows the measured power factor over the range of universal line input at full load, in which the maximum power factor approaches unity. Figs. 21 and 22 illustrate the prototype efficiency. Fig. 21 shows the efficiency curve from 4 W to 100 W, while line voltage is 110 Vrms. The figure also demonstrates that DOSSBS can achieve the highest efficiency among the converters of conventional full-bridge SEPIC PFC, bridgeless SEPIC PFC, and bridgeless non-SEPIC PFC. The maximum efficiency of the prototype is up to 95% at approximately 60 W. Efficiency measurement over the entire range of universal line input under full load is presented in Fig. 22, in which the efficiencies at 110 and 220 Vrms are 93.5% and 95.7% respectively. A comparison with other types of single-stage PFC is summarized in Table II. The proposed converter does not require low-speed diode and dual-output topology.

Fig. 10.Measured waveforms of the line voltage vs and the input inductor current iL1 at full load.

Fig. 11.Zoomed-in waveforms of input inductor current and associated control signal in the positive half-line cycle.

Fig. 12.Measured waveforms of line voltage and filtered source current.

Fig. 13.Measured waveforms of output voltages at ports 1 and 2.

Fig. 14.Related waveforms while the output power of port 1 changes from light to heavy load.

Fig. 15.Related waveforms while the output power of port 1 changes from heavy to light load.

Fig. 16.Measured waveforms of vC1 and vC2.

Fig. 17.Measured waveforms of vD1 and vD2.

Fig. 18.Measured waveforms of vD3 and vD4.

Fig. 19.Measured result of current harmonics.

Fig. 20.Measured power factor over the range of universal line input at full load.

Fig. 21.Efficiency measurement from light to full load while line voltage is 110 Vrms.

Fig. 22.Efficiency measurement over the range of universal line input from 85 Vrms to 265 Vrms.

TABLE IICOMPARISON AMONG THE PROPOSED CONVERTER AND OTHER TYPES OF SINGLE -STAGE PFC

 

V. CONCLUSION

This study proposes DOSSBS PFC, which can deal with a wide range of input of 85-265 Vrms of universal line and provide dual outputs. In the proposed converter, the front-end rectifier is completely removed, thereby simplifying configuration, decreasing component count, and reducing conduction losses. A coupled inductor is incorporated to replace two separate inductors and thus reduce converter volume, as well as recycle the energy stored in leakage inductance. Practical measurements validate the proposed DOSSBS, whose configuration can be expanded for multiple-output applications. Fig. 23 shows the main power schematics, in which the inductors of all the additional output ports can be coupled with Lpri and Lsec.

Fig. 23.Main circuit schematic of the expanded configuration of the proposed converter.