A Three-Phase AC-DC High Step-up Converter for Microscale Wind-power Generation Systems

Abstract

In this paper, a three-phase AC-DC high step-up converter is developed for application to microscale wind-power generation systems. Such an AC-DC boost converter prossessess the property of the single-switch high step-up DC-DC structure. For power factor correction, the advanced half-stage converter is operated under the discontinuous conduction mode (DCM). Simulatanously, to achieve a high step-up voltage gain, the back half-stage functions in the continuous conduction mode (CCM). A high voltage gain can be obtained by use of an output-capacitor mass and a coupled inductor. Compared to the output voltage, the voltage stress is decreased on the switch. To lessen the conducting losses, a low rated voltage and small conductive resistance MOSFETs are adopted. In addition, the coupled inductor retrieves the leakage-inductor energy. The operation principle and steady-state behavior are analyzed, and a prototype hardware circuit is realized to verify the performance of the proposed converter.

I. INTRODUCTION

Since the world population is rapidly growing, the demand for energy is rising and fossil fuels, including oil, gas, and coal are gradually being depleted. In addition, utilizing fossil fuels has resulted in a rapid accumulation of carbon dioxide in the past few decades. This has resulted in greenhouse effects and abnormal global climate changes. In an effort to solve this energy shortage and to reduce pollution, sustainable power sources, namely photovoltaic, wind-power, geothermal, hydraulic, hydro, biomass, etc., are being gradually developed [1]-[5]. In wind energy generating systems, the output-voltage and frequency of the generator are varied due to wind speed. Thus, their ranges are very large. Traditionally, a bridge-diode rectifier in combination with a DC-DC structure is adopted in converting AC to DC power for wind energy generating systems, as shown in Fig. 1(a) [6]-[8]. This results in a large pulsating input current and large current stresses in the generator windings. To resolve these difficulties, various circuits have been presented for power factor correction, including buck type [9]-[11], boost type [12]-[14], and buck-boost type [15]-[18]. These converters can be operated in the continuous conduction mode (CCM) or the discontinuous conduction mode (DCM). As a result, a higher power factor and lower current stresses for the generator can be achieved. For wind-power generation systems, grid-connection is the main application. For the purpose of boosting the output-voltage in a generator, a high DC voltage must be produced, as shown in Fig. 1(b). Although Boost converters can be applied for the conversion of step-up voltage, a high voltage gain cannot be provided, which has led to the study of high gain step-up converters [19]-[24].

Fig. 1.Wind-power generation system. (a) Diode-bridge rectifier type, (b) Power factor correction type.

A single-stage three-phase AC-DC converter with a high step-up structure was investigated for micro-scale wind-power generation systems, as shown in Fig. 2(a). Some of the part contents have been presented in the literature [25]. This type of converter utilizes a single switch to integrate a conventional three-phase AC-DC boost converter and a high step-up DC-DC structure with a coupled inductor [26]-[30]. The presented converter yields a high power factor, which alleviates the large pulsating current and reducing the current stresses of the generator windings. For the sake of power factor correction, the advanced half-stage circuitry, i.e., a three-phase AC-DC boost converter is fulfilled in the DCM while the back half-stage associated with the output-capacitor stack and coupled-inductor techniques works in the CCM for a high gain step-up voltage. In addition, the active-clamp function is achieved and smaller switch voltage stress yields are obtained when compared with the output-voltage. It is worth mentioning that the decreased losses in conducting can be obtained by low rated voltage and small conductive resistance MOSFETs. In addition, the leakage-inductor energy of the coupled inductor is also regained.

Fig. 2.Proposed converter. (a) Circuit configuration, (b) Simplified circuit model.

 

II. OPERATING PRINCIPLES

Fig. 2(b) reveals a simplified circuitry model of the proposed converter. The coupled inductor can be represented by a magnetizing inductor Lm, a primary leakage inductor Lk1, a secondary leakage inductor Lk2, and an ideal transformer. To analyze the circuitry simply, the following circumstances are supposed: 1) Allow all of the elements to be ideal. That is, omit the ON-state resistance Rds(on) of the active switch, the forward voltage drop of the diodes, and the equivalent series resistance of the coupled inductor and output capacitors. 2) When the output capacitors C1, C2, and C3 are sufficiently large, the voltages across these capacitors can be regarded as constant at the each switching duration.

The pulse-width-modulation technology is applied for controlling the switch S1. The advanced half-stage of the proposed converter is performed under the DCM. The back half-stage of the proposed converter is fulfilled under the CCM. The inductances of the three input inductors are equal, namely La = Lb = Lc = L. Because a three-phase system is symmetrical, the principle of operation is explored within [0°, 30°]. Several classic waveforms at a duration of one switching can be illustrated as in Fig. 3.

Fig. 3.Some waveforms in one switching period during [0°, 30°].

(1) Mode 1, [hTs, th1]: Fig. 4(a) shows the direction of the current-flow. When the switch S1 is turned on, the energies of the three input inductors La, Lb, and Lc are reserved through line source. The absolute values of the currents iLa, iLb, and iLc raise, and the sum of the absolute values of the currents iLa and iLc agree with the absolute value of the current iLb. The capacitor C1 delivers energy to the magnetizing inductor Lm and primary leakage inductor Lk1 owned by coupled inductor. The secondary leakage inductor Lk2 owned by the coupled inductor retrieves energy to the capacitor C3. Therefore, the magnetizing-inductor current iLm and primary leakage-inductor current iLk1 are raised and the secondary leakage-inductor current iLk2 is reduced. The stacked energies from the capacitors C1, C2, and C3 can provide the load. At t = tk1, the energy storage by the secondary leakage inductor Lk2 is empty. The current iLk1 is equal to the current iLm.

Fig. 4.Current-flow path of proposed converter during [0°, 30°].

(2) Mode 2, [th1, th2]: Fig. 4(b) shows the direction of the current-flow, and the switch S1 turns on. The energies of the three input inductors La, Lb, and Lc are continuously derived via the line source. Therefore, the absolute values of the currents iLa, iLb, and iLc, still raise, and the sum of the absolute values of the currents iLa and iLc agrees with the absolute value of the current iLb. The capacitor C1 still delivers energy to the magnetizing inductor Lm and primary leakage inductor Lk1. The currents iLm and iLk1 are increased, and the stacked energies from the capacitors C1, C2, and C3 are provided to the load.

(3) Mode 3, [th2, th3]: Fig. 4(c) indicates the direction of the current-flow, and the switch S1 turns off. The three input inductors La, Lb, and Lc deliver energies to C1. Therefore, the absolute values of the currents iLa, iLb, and iLc are reduced, and the sum of the absolute values of the currents iLa and iLc agrees with the absolute value of the current iLb. The magnetizing inductor Lm and primary leakage inductor Lk1 owned by the coupled inductor deliver energies to the capacitor C2, and the magnetizing inductor Lm discharges part of the energy to the secondary leakage inductor Lk2 and capacitor C3. Therefore, the currents iLm and iLk1 are decreased and the current iLk2 is increased. The primary leakage inductor Lk1 retrieves the energy from the capacitor C2. The stacked energies from the capacitors C1, C2, and C3 are provided to the load. At t = tk3, there is a blank energy storage by the inductor La.

(4) Mode 4, [th3, th4]: Fig. 4(d) reveals the direction of the current-flow, and the switch S1 turns off. The inductors Lb and Lc continuously deliver energies to the capacitor C1. Therefore, the absolute values of the currents iLb and iLc reduce, and the absolute value of the current iLb agrees with the absolute value of the current iLc. The magnetizing inductor Lm and the primary leakage inductor Lk1 deliver energies to the capacitor C2, and the magnetizing inductor Lm discharges part of the energy to the secondary leakage inductor Lk2 and the capacitor C3. Therefore, the currents iLm and iLk1 are decreased, and the current iLk2 is increased. Meanwhile, the primary leakage inductor Lk1 retrieves energy from the capacitor C2. The stacked energies from the capacitors C1, C2, and C3 are provided to the load. At t = th4, the energy storage by the primary leakage inductor Lk1 is empty.

(5) Mode 5, [th4, th5]: Fig. 4(e) shows the direction of the current-flow, and the switch S1 turns off. The inductors Lb and Lc deliver energies to the capacitor C1. Therefore, the absolute values of the currents iLb and iLc are reduced and the absolute value of the current iLb agrees with the absolute value of the current iLc. The magnetizing inductor Lm and secondary leakage inductor Lk2 discharge energies to the capacitor C3. Therefore, the currents iLm and iLk2 are decreased. Meanwhile, the secondary leakage inductor Lk2 retrieves the energy from the capacitor C3. The stacked energies from the capacitors C1, C2, and C3 are provided to the load. At t = th5, there are blank energy storages by the inductors Lb and Lc.

(6) Mode 6, [th5, (h+1)Ts]: Fig. 4(f) denotes the direction of the current-flow, and the switch S1 turns off. The magnetizing inductor Lm and secondary leakage inductor Lk2 discharge energies to the capacitor C3. Thus, the currents iLm and iLk2 are decreased. Meanwhile, the secondary leakage inductor Lk2 retrieves the energy from the capacitor C3. The stacked energies from the capacitors C1, C2, and C3 are provided to the load.

 

III. STEADY-STATE ANALYSIS

A. Front Semi-Stage of the Proposed Converter

Because the three-phase system provides a symmetrization property, the steady-state behavior is analyzed below for the duration of [0°, 30°]. Briefly, omit the influence caused by the input filter and assume that the three input phase voltages are:

where Vm denotes the amplitude of the input phase voltage.

Since the switching frequency fs is far bigger than the line frequency f1, the input phase voltages can be regarded as a piecewise constant for the duration of each switching period. Let the parameter m be the switching number at the duration of [0°, 30°]. Then the parameter m is equal to fs/12f1. The analysis in the following is investigated within the switching period [hTs, (h+1)Ts], where h = 0, 1,..., m-1. While the switch S1 is turned on, the equations yields:

Thus, the currents iLa, iLb, and iLc are obtained as follows:

At t = th2, the peak values of iLa, iLb, and iLc are found to be:

where D indicates the duty ratio.

Within [th2, (h+1)Ts], the switch S1 turns off. During [th2, th3], the formulas can be obtained by Fig. 4(c).

Through the use of (5), the voltages across the inductors La, Lb, and Lc are derived as:

From Fig. 3, the voltages vLa, vLb, and vLc can also be determined as follows:

where tr1,h = th3 - th2.

Substituting (4) and (6) into (7), the interval tr1,h is obtained as:

(7) can be replaced by (4), (6) and (8), and then it is possible to obtain:

During [th3, th5], it is possible to obtain the formulas below by means of Figs. 4(d) and 4(e).

Thus:

where tr2,h = th5 - th3.

Substituting (9) into (11), yields:

B. Rear Semi-Stage of the Proposed Converter

While the switch S1 turns, the voltage across the magnetizing inductor Lm can be derived as:

where the coupling coefficient k of the coupled-inductor agrees with Lm/(Lm+Lk1). Therefore:

While the switch S1 turns off, the following equation is obtained:

where the turns ratio n of the coupled inductor agrees with N2/N1. Owing to k = Lm/(Lm+Lk1):

Thus:

Substituting (16) into (18), yields:

Substituting (19) into (15), yields:

Therefore:

By the principle of the volt-second balance acted on by the magnetizing inductor Lm, it is possible to obtain:

(21) is replaced by (13) and (19). Then the equations become:

From the operating principle, it is known that Lk1 retrieves energy from the capacitor C1. By the principle of the ampere-second balance on C1, the voltage across C1 can be obtained as follows [31]:

Then:

At k = 1, equation (25) can be rewritten as:

Therefore, the proposed converter can provide a high voltage gain.

 

IV. EXPERIMENTAL RESULTS

A laboratory prototype is realized to demonstrate the effectiveness of the proposed converter for micro-scale wind-power generation systems. The output voltage and frequency produced by the three-phase generator match the input voltage and frequency created by the proposed converter. The system parameters of the proposed converter are chosen as: input voltage/frequency Vin/f1 = 20Vrms/15Hz ~ 80Vrms/62Hz, output voltage Vo = 400 V, output power Po = 600 W, switching frequency fs = 50 kHz, input inductor L = 34 μH, coupled inductor Lm = 205 μH (turns ratio n = 1), capacitors C1 = 820 μF, C2 = C3 = 220 μF, input filter Lf = 2.4 mH, Cf = 2 μF, switch S1: IXFK64N50P, and diodes D1~D4: DSEP30-06A.

Fig. 5 shows a circuit diagram of the proposed converter with its control circuitry. Figs. 6 and 7 show some experimental waveforms under full-load conditions, Vin/f1 = 80 Vrms/62 Hz and Po = 600 W, and light-load conditions, Vin/f1 = 20 Vrms/15 Hz and Po = 30 W. It can be seen from Figs. 6(a) and 7(a) that the input phase voltage and input phase current are in phase. Figs. 6(b) and 7(b) show the waveforms of the three input currents. It can be seen that the current stress is reduced for the generator winding. The waveforms of the input inductor currents iLa, iLb, and iLc are represented in Figs. 6(c) and 7(c). It can be seen that the advanced half-stage is performed under the DCM. Figs. 6(d) and 7(d) show the waveforms of the leakage-inductor currents iLk1 and iLk2. Waveforms of the three output-capacitor voltages are plotted in Figs. 6(e) and 7(e). It can be seen that the sum of Vc1, Vc2, and Vc3 agrees with the output voltage. Fig. 8 shows the efficiency of the measurements under various conditions. The maximal efficiency is 88% at Vin/f1 = 61 Vrms/46 Hz, Vo = 400 V, and Po = 360 W.

Fig. 5.Proposed converter with control circuit.

Fig. 6.Some experimental waveforms at full-load conditions Vin/f1 = 80 Vrms/62 Hz, Vo = 400 V, and Po = 600 W. (a) Input phase voltage and input phase current. (b) Three input phase currents. (c) Three input inductor currents. (d) Leakage-inductor currents iLk1 and iLk2. (e) Three output-capacitor voltages.

Fig. 7.Some experimental waveforms at light-load conditions Vin/f1 = 20 Vrms/15 Hz, Vo = 400 V, and Po = 30 W. (a) Input phase voltage and input phase current. (b) Three input phase currents. (c) Three input inductor currents. (d) Leakage-inductor currents iLk1 and iLk2. (e) Three output-capacitor voltages.

Fig. 8.Measured efficiency at Vin/f1 = 20 Vrms/15 Hz~80 Vrms/62 Hz, Vo = 400 V, and Po = 30~600 W.

 

V. CONCLUSIONS

A three-phase high step-up AC-DC converter is researched for micro-scale wind power generation systems. The advanced half-stage of the proposed converter can provide a high power factor. Output-capacitor stack and coupled inductor technologies are used for the back half-stage of the proposed converter to achieve a high voltage gain. Additionally, the energy of the leakage-inductor can be retrieved. A laboratory prototype is realized to confirm the theoretical analysis. From the experimental results, it can be seen that a high power factor and a high voltage gain can be obtained for different input line voltages/frequencies and output powers.