I. INTRODUCTION
DC–DC converters have been widely applied in the industrial applications, such as in hybrid electric vehicles and renewable energy systems [1], [2]. To attain high density and efficiency, soft-switching techniques are utilized in high-frequency DC–DC converters. Among these techniques, zero-voltage-switching (ZVS) [3]-[26] including quasi-resonant ZVS, active-clamping ZVS, zero-voltage-transition (ZVT), and magnetic coupling ZVS converters have been intensively studied to eliminate turn-on losses, which is the dominant switching losses in the majority carrier devices, such as MOSFET.
Quasi-resonant converters (QRCs) achieve ZVS operation by employing LC resonance to create a zero-voltage turn-on condition [3]-[5]. Switching losses are effectively reduced, but the main switch suffers from excessively high voltage stress because of resonance. Moreover, the switching frequency varies widely under large load variations, thus making the design of passive components difficult. Pulse-width modulation (PWM) can be achieved in active-clamping ZVS converters with the assistance of an auxiliary switch [6]-[10]. However, the circulating current in the auxiliary circuit is large, thus increasing conduction losses. In addition, the voltage stress of the main switch remains high.
Conduction losses are reduced in ZVT converters since the auxiliary circuit is removed from the main circuit and only activated in a short interval around the turn-on of the main switch [11]-[21]. Furthermore, the voltage stress of the main switch is lower than that of QRCs and active-clamping converters. The work in [11] classified ZVT converters into three classes according to the implementation of the auxiliary voltage source (AVS): Class-A with switched AVS [12]-[15], Class-B with DC AVS [16]-[18], and Class-C with resonant AVS [19]-[21]. The characteristics of the main circuit in all these three converters are similar, whereas the performances of their auxiliary circuits are quite different [22]. The auxiliary circuit of Class-A ZVT converters is simple, but their auxiliary switch is zero-current-switching (ZCS) turned on and ZVS turned off, which causes additional switching losses. Improved ZCS turn-on and turn-off is achieved for the auxiliary switch in the Class-B and Class-C ZVT converters. But in Class-C ZVT converters with resonant AVS, the design of the resonant tank is complex, and additional reactive energy is introduced. In Class-B ZVT converters, the configuration of the auxiliary circuit remains simple because the DC AVS can be directly obtained from the voltage source of the main circuit. However, the available input voltage range and load variation are limited to ensure that ZVS operation is achieved for the main switch [16], [17].
In references [23]-[26], coupled inductors were employed to achieve ZVS for synchronous DC–DC converters over a wide range of input voltage and load variation. The auxiliary circuit is relatively simple because it is composed of an auxiliary diode and a coupled winding. The magnetic component is reduced since the leakage inductor of the coupled inductor is utilized to function as the auxiliary inductor, which is indispensable in non-isolated QRCs, active-clamping ZVS converters, and ZVT converters. However, the circulating conduction loss in the auxiliary circuit is large because the auxiliary circuit is uncontrolled and activated in a long interval. In [27], circulating conduction losses were reduced with the addition of a switch in series with the auxiliary diode, and the auxiliary circuit was only activated in a short interval. Meanwhile, the controlled auxiliary circuit was also utilized in [28] to achieve ZVS operation for bidirectional DC–DC converters.
The present study proposes a family of magnetic coupling ZVS DC–DC converters. A pair of auxiliary switch and diode is utilized to control the auxiliary circuit, thereby contributing to small circulating conduction losses. Besides, with the assistance of the coupled winding, soft-switching is achieved for both main and auxiliary switches over wide input voltage range and load variation, along with a reduced magnetic component. The paper is organized as follows. Class-B ZVT DC–DC converters are briefly introduced, and a family of magnetic coupling ZVS buck, boost as well as buck-boost converters is derived in Section II. As an example, the operation principle of the proposed ZVS buck converter is shown in Section III. The analysis and comparison results with other ZVS buck converters are provided in Section IV. The experimental results validating the effectiveness of the proposed converter are presented in Section V. Finally, conclusions are drawn in Section VI.
II. TOPOLOGY DERIVATION
In order to gain a clear understanding of the soft-switching principle, Class-B ZVT DC-DC converters which implement DC AVS without additional components, are briefly introduced. ZVS and ZCS can be respectively achieved for main and auxiliary switches with a simple auxiliary circuit. However, ZVS realization is restricted by the input voltage and load condition. To improve the soft-switching characteristic, a family of magnetic coupling DC–DC converters is proposed, which realizes ZVS for the main switch over wide input voltage range as well as load variation. Also, the benefit of ZCS for the auxiliary switch is maintained. Furthermore, no extra auxiliary inductor is needed, contributing to reduced magnetic component.
A. Class-B ZVT DC–DC Converters
The general Class-B ZVT DC–DC converter composed of a basic PWM block and a simple auxiliary circuit which implements DC AVS without additional components, is illustrated in Fig. 1(a) [16]-[18]. The basic PWM block consists of the inductor L1, main diode D1 with parasitic capacitor Cd, and main switch S1 with parasitic capacitor Cds. To achieve ZVS for the main switch S1, the auxiliary circuit makes use of the inductor La, switch Sa, and diode Da to discharge and charge Cds and Cd, respectively. The operation of the converter is the same as that of the conventional DC–DC converter, except when the auxiliary circuit is active. In the active interval, key waveforms including the drive signals of the main and auxiliary switches, drain-to-source voltage vds, as well as auxiliary inductor current iLa, are shown in Fig. 1(b) [11], and the equivalent circuit is illustrated in Fig. 1(c). To simplify the operation principle, the current of inductor L1 is assumed to be constant and denoted as IL.
Fig. 1.Class-B ZVT DC-DC converter: (a) general topology, (b) key waveforms, and (c) equivalent circuit in the interval t0–t4.
Before t0, the auxiliary switch Sa is in the off state, and the auxiliary inductor current iLa is zero. The switch S1 is in the off state, and the inductor current IL flows through the diode D1.
Stage 1 (t0–t1): At t0, the auxiliary switch Sa is turned on. Then the auxiliary inductor La is charged, and iLa is linearly increased to IL at t1. The interval t1–t0 is obtained in (2).
Stage 2 (t1–t2): In this stage, the equivalent capacitance Ceq, which is the sum of parasitic capacitances Cds and Cd, resonates with the auxiliary inductor La, as illustrated in (3). Then the voltage vds and current iLa can be derived in (4). At t2, the voltage vds decreases to zero, and the resonant process is finished. The interval t2–t1 and iLa(t2) are given in (5) and (6), respectively.
where .
Stage 3 (t2–t3): After the complement of the resonant process, the current iLa-IL flows through the anti-parallel diode of switch S1. Therefore, iLa is linearly decreased, and the switch S1 should be turned on to achieve ZVS operation before iLa decays to IL at t3. Thus, the phase angle φ between the two drive signals in Fig. 1(b) should satisfy (9).
where Ts is the switching period.
Stage 4 (t3–t4): The operation in this stage is similar with that of stage 3, except that the current iLa-IL flows from the anti-parallel diode to the MOSFET channel of switch S1. iLa continues to decrease and drops to zero at t4.
In order to achieve ZVS for the main switch S1, the parasitic capacitance Cds should be completely discharged in stage 2; thus, the voltage limitation is derived in (10) from (4) [16]. Meanwhile, the limit of maximum inductor current IL,max is obtained in (11) from (2), (5), (8), and (9) to realize ZVS with a fixed φ over the whole load range. Therefore, the practical applications of Class-B ZVT converters in Fig. 1(a) are restricted.
B. Proposed Magnetic Coupling ZVS DC–DC Converters
In order to achieve ZVS for main switches over wide input voltage range and load variation, a family of magnetic coupling ZVS DC-DC converters is proposed. The general magnetic coupling ZVS DC–DC converter is illustrated in Fig. 2(a). Compared with the Class-B ZVT converter in Fig. 1(a), a coupled inductor T1 with turns ratio Np:Ns=1:n is employed to implement the main inductor L1 with the magnetizing inductor Lm and the auxiliary inductor La with the leakage inductor Lr. Thus the magnetic component number is reduced. The secondary winding of the coupled inductor T1, the auxiliary switch Sa, and the diode Da are connected in series between ports c and d. The equivalent circuit in the interval when Sa and Da conduct is shown in Fig. 2(b). Compared with Fig. 1(c), an additional voltage source -Vcd/n is added since the secondary voltage of the coupled inductor T1 is clamped to -Vcd. Hence, the limitation in (10) and (11) is modified in (12) and (13), respectively. According to (10), for the Class-B ZVT converter in Fig. 1(a), the ZVS operation of the main switch S1 can be achieved only when Vcd is smaller than 0.5 Vad, which is greatly alleviated in the proposed converter as illustrated in (12). Moreover, the available ZVS load range is also enlarged with the comparison of (13) and (11).
Fig. 2.Proposed general magnetic coupling ZVS DC–DC converter: (a) topology configuration and (b) equivalent circuit in the active interval.
On the basis of the general magnetic coupling DC–DC converter in Fig. 2(a), the corresponding ZVS buck, boost, and buck-boost converters with different values for Vad and Vcd in the Table I, are derived in Fig. 3. Besides, the magnetic coupling concept can be employed to achieve ZVS operation for various converters which consists of the basic PWM block.
TABLE IVALUES OF VAD AND VCD FOR PROPOSED ZVS BUCK, BOOST, AND BUCK-BOOST CONVERTERS
Fig. 3.Proposed magnetic coupling ZVS DC–DC converters: (a) buck, (b) boost, and (c) buck-boost.
III. OPERATION PRINCIPLE
As the operation principles of the proposed magnetic coupling ZVS buck, boost, and buck-boost converters are similar, the ZVS buck converter is used as an example in this section, which is re-shown in Fig. 4(a). Key operating waveforms of the converter are shown in Fig. 4(b). The auxiliary switch Sa is turned on in advance to decrease the leakage inductor current iLr and create ZVS condition for the main switch S1. In a switching period, the operation comprises seven stages, and the corresponding equivalent circuits of which are illustrated in Fig. 5.
Fig. 4.Proposed magnetic coupling ZVS buck converter: (a) circuit and (b) key waveforms.
Fig. 5.Equivalent circuits of the magnetic coupling ZVS buck converter: (a) stage 1, (b) stage 2, (c) stage 3, (d) stage 4, (e) stage 5, (f) stage 6, and (g) stage 7.
Several assumptions are made to simplify the operation principle.
From Fig. 4(a), the relationship among the leakage inductor current iLr, auxiliary current ia, magnetizing current IL, and output current io is derived in (14) and (15).
Prior to t0, S1 and Sa are in the on state. The leakage inductor current iLr is increased, whereas the auxiliary current ia is decreased.
Stage 1 (t0–t1): At t0, iLr rises to the magnetizing current IL, and ia decays to zero; thus, Da is reverse biased. In this stage, the energy is transferred from the input to the leakage inductor Lr, magnetizing inductor Lm, and output. Since the auxiliary current ia is zero, the auxiliary switch Sa is ZCS turned off in this stage.
Stage 2 (t1–t2): S1 is turned off at t1. The parasitic capacitors Cds and Cd are respectively charged and discharged by IL. Therefore, the drain-to-source voltage vds increases linearly, as shown in (19).
Stage 3 (t2−t3): At t2, vds increases to Vi; hence, the main switch D1 is forward biased. During this stage, the energy is transferred from the leakage inductor Lr and magnetizing inductor Lm to the output.
Stage 4 (t3–t4): At t3, the auxiliary switch Sa is turned on, and the magnetizing voltage vLm is clamped to (Vi−Vo)/n. In this stage, the leakage inductor current iLr and the diode current iD decrease linearly, as shown in (27). Based on (14), the auxiliary current ia increases.
Stage 5 (t4–t5): The leakage inductor current iLr and diode current iD decay to zero at t4. Owing to the leakage inductor Lr, the reverse-recovery problem of the main diode is greatly alleviated. Then, Lr resonates with Ceq as shown in (28).
Stage 6 (t5–t6): The voltage vds decreases to zero at t5, and the current iLr flows through the body diode of S1; hence, S1 is ZVS turned on. The leakage inductor voltage vLr, shown in (29), must be larger than zero to increase the leakage inductor current iLr to the magnetizing current IL. Therefore, the turns ratio n should be larger than 1. And from (14), the auxiliary current ia decreases.
Stage 7 (t6–t7): At t6, iLr rises to zero. This stage is similar with the stage 6 except that the leakage inductor current iLr is positive. The switching period ends at t7 when the auxiliary current ia drops to zero.
IV. ANALYSIS AND COMPARISON
A. Voltage Gain
Owing to the flux balance, the average voltage across the leakage inductor Lr and magnetizing inductor Lm in a switching period is zero. Therefore, the output voltage Vo is equal to the average diode voltage VD. From Fig. 4(b), the voltage gain of the proposed converter is derived in (32), which is similar with that of the conventional buck converter.
B. Average Magnetizing Current IL
From the operation principle in section III, the leakage inductor current iLr in different stage is summarized in (33). Then the relationship between the average leakage inductor current ILr and magnetizing current IL can be derived. Combing with (34) derived from (14) and (15), IL can be calculated in terms of average output current Io. The relationship between IL and Io with Lr=6 μH and n=1.5 as a function of voltage gain M is illustrated in
Fig. 6. The average magnetizing current IL is a little larger than the average output current Io, but the difference is small, particularly at a low voltage gain M.
Fig. 6.Relationship between IL and Io as a function of voltage gain M.
C. Voltage Stress
The turn-off voltage of the main switch S1 and diode D1 is clamped by the input voltage Vi, which is the same as the conventional buck converter. In the stage 1, the auxiliary diode Da is reverse biased and the turn-off voltage is given in (36) with the neglect of the leakage inductance Lr, which is much smaller than the magnetizing inductance Lm. Likewise, the auxiliary switch Sa is turned off in the stage 3, and the voltage stress is derived in (37).
D. Current Stress
According to Fig. 4(b), in the intervals [t5, t7] and [t0, t1], the main switch S1 conducts and the current iS is equal to the leakage inductor current iLr. Therefore, the RMS current of the main switch is derived in (38). In the interval [t2, t4], the leakage inductor current iLr flows through the main diode D1 and hence the average current ID1 is given in (39). Similarly, the RMS current of the auxiliary switch Ia,RMS and the average current of the auxiliary diode Ia,ave can be obtained, as illustrated in (40) and (41).
E. Soft-switching Analysis
For the proposed buck converter, in order to achieve ZVS for the main switch S1, (42) must be satisfied from (12) and Table I. Combining with n>1 in (30), the available value of turns ratio n at different voltage gain M is depicted in Fig. 7. With 1 < n ≤ 2, the main switch can theoretically realize ZVS over the whole input voltage range.
Fig. 7.Available turns ratio n to achieve ZVS as a function of M.
In practical, the maximum current IL,max is restricted in (11) and (13) to realize ZVS operation over the whole load range with a fixed phase angle φ for the Class-B ZVT buck converter in Fig. 1(a) and the proposed converter, respectively. Fig. 8 depicts the relationship between IL,max and the voltage gain M with different turns ratios n. The available ZVS load range for the ZVT buck converter in Fig. 1(a) is narrow and deteriorated with the increase of auxiliary inductance La, which is significantly improved in the proposed ZVS buck converter. Moreover, with the decrease of turns ratio n, the available load range is enlarged. Therefore, the proposed buck converter can achieve ZVS operation for the main switch over both wide input voltage range and large load variation.
Fig. 8.Maximum magnetizing current IL,max to achieve ZVS over whole load range with a fixed φ for different converters.
The decreasing ratio of the diode current iD and auxiliary current ia is limited by the leakage inductor Lr, as illustrated in Fig. 4(b). Therefore, the reverse-recovery problem of diodes D1 and Da is greatly alleviated. Moreover, the auxiliary switch Sa is ZCS turned off in the stage 1 since the auxiliary current ia already decays to zero at t0.
F. Comparison
The comparison results between the proposed and other ZVS buck converters are shown in Table III. The QRC buck converter in [4] is considerably simple and does not require an auxiliary switch, but the voltage stress of the main switch is excessively high. In [6], the voltage stress was reduced in the active-clamping buck converter, but it remained higher than that of the input voltage Vi. Moreover, the circulating current is large; thus, conduction losses are deteriorated. The ZVT buck converter in [17] can effectively reduce conduction losses because the auxiliary circuit is removed from the main circuit and only activated in a short interval. However, ZVS operation is lost in the application with Vo<0.5Vi. Nevertheless, all aforementioned converters need an auxiliary inductor. In the proposed buck converter, the ZVS operation can be achieved over wide input voltage range and load variation with the benefits of low voltage stress on the main switch, small circulating current and reduced magnetic component.
V. EXPERIMENTAL VALIDATION
In order to verify the effectiveness of the proposed topology, a 500 W prototype circuit of the magnetic coupling ZVS buck converter with topology parameters in Table II is built. The soft-switching characteristics of the converter are assessed at different input voltages 200 and 300 V under 10% and 100% load variations with open-loop experimental results, respectively.
TABLE IIPARAMETER SPECIFICATION
TABLE IIICOMPARISON OF THE PROPOSED ZVS BUCK CONVERTER AND OTHER ZVS BUCK CONVERTERS
The steady-state experiment waveforms at 100% load with input voltages Vi=200 V and Vi=300 V are shown in Fig. 9, Fig. 10 and Fig. 11, which are in well coincidence with the theoretical analysis. In Fig. 9, the leakage inductor current iLr decreases after the turn-on of auxiliary switch Sa to achieve ZVS turn-on for the main switch S1, and then rapidly reset to the magnetizing current IL. The auxiliary circuit is only activated in a short interval, and thus the additional conduction loss is reduced. It is noteworthy that the auxiliary current ia increases slightly and that the leakage inductor current iLr decreases after the turn-off of the main switch S1 due to the influence of the RCD, which is in parallel with the auxiliary switch Sa. From Fig. 10, ZVS is achieved for the main switch S1 at both input voltages. Moreover, the reverse-recovery problem of the main diode D1 is greatly alleviated. In addition, as shown in Fig. 11, the ZCS of the auxiliary switch Sa is obtained, which is also independent of the input voltage. Therefore, the proposed ZVS buck converter achieves desirable soft-switching characteristics over a wide input voltage range.
Fig. 9.Experiment waveforms of drive signals vgs1–vgsa, leakage inductor current iLr, and auxiliary current ia, at 100% load with different input voltages: (a) Vi=200 V and (b) Vi=300 V.
Fig. 10.Experiment waveforms of drive signals vgs1, drain-to-source voltage vds, current iS, and diode current iD at 100% load with different input voltages: (a) Vi=200 V and (b) Vi=300 V.
Fig. 11.Experiment waveforms of drive signals vgsa, drain-to-source voltage vsa, and current ia at 100% load with different input voltages: (a) Vi=200 V and (b) Vi=300 V.
The experiment waveforms of the main switch S1 and auxiliary switch Sa at 10% load condition with input voltages Vi=200 V and Vi=300 V are illustrated in Fig. 12 and Fig. 13, respectively. ZVS and ZCS operation are also achieved. Note that the resonance caused by the non-ideality of semiconductor devices and parasitic inductances during the turn-on process of S1 has slight influence on the ZVS operation, e.g. the drain-to-source voltage vds increases slightly as shown in Fig. 12(b). Nevertheless, vds finally decays to zero before the turn-on of S1. Therefore, improved soft-switching characteristics of the proposed converter remain regardless of the load variation.
Fig. 12.Experiment waveforms of the drive signal vgs1, drain-to-source voltage vds, and current iS at 10% load with different input voltages: (a) Vi=200 V and (b) Vi=300 V.
Fig. 13.Experiment waveforms of the drive signal vgsa, drain-to-source voltage vsa, and current ia at 10% load with different input voltages: (a) Vi=200 V and (b) Vi=300 V.
The measured efficiency of the proposed magnetic coupling buck converter at Vi=200 V and Vi =300 V is shown in Fig. 14(a), which achieves a maximum value of 97.2%. Thanks to the desirable soft-switching characteristic and the small auxiliary circuit conduction loss, the efficiency of the proposed converter is much improved over whole load range at both input voltages, in comparison with the conventional buck converter, as illustrated in Fig. 14(b)-(c). Compared with the ZVT buck converter in Fig. 1(a), the proposed converter achieves a slightly lower efficiency under heavy load with input voltage Vi=200 V in Fig. 14(b). Nevertheless, the efficiency of the ZVT converter under light load is harshly decreased, resulting from the loss of ZVS with a fixed phase angle φ. Moreover, at Vi=300 V in Fig. 14(c), ZVS operation is still obtained in the proposed ZVS buck converter but lost in the ZVT buck converter. Therefore, the efficiency of the proposed converter is higher than that of the ZVT converter at Vi=300 V. A photograph of the prototype circuit is shown in Fig. 15.
Fig. 14.Measured efficiency: (a) proposed converter at Vi=200 V and Vi=300 V, (b) comparison at Vi =200 V, and (c) comparison at Vi =300 V.
Fig. 15.Photograph of the prototype circuit.
VI. CONCLUSIONS
A family of magnetic coupling ZVS buck, boost and buck-boost converters is proposed. Thanks to the additional auxiliary voltage source provided by the coupled inductor, ZVS operation is achieved for the main switch over wide input voltage range and load variation with a reduced magnetic component. Besides, the auxiliary circuit obtains the advantages of ZCS operation and small circulating current. The proposed ZVS buck converter is taken as an example to clearly introduce the operation principle and analysis. Experiment waveforms and measured efficiency based on a 500 W prototype circuit at input voltages of 200 V and 300 V under 10% and 100% load conditions are given to validate converter performance.
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