Journal of Power Electronics

The Korean Institute of Power Electronics (전력전자학회)
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Long-Lasting and Highly Efficient TRIAC Dimming LED Driver with a Variable Switched Capacitor

  • Lee, Eun-Soo (Department of Nuclear and Quantum Engineering, KAIST) ;
  • Choi, Bo-Hwan (Department of Nuclear and Quantum Engineering, KAIST) ;
  • Nguyen, Duy Tan (Department of Nuclear and Quantum Engineering, KAIST) ;
  • Choi, Byeung-Guk (Department of Nuclear and Quantum Engineering, KAIST) ;
  • Rim, Chun-Taek (Department of Nuclear and Quantum Engineering, KAIST)
  • Received : 2015.11.16
  • Accepted : 2016.02.19
  • Published : 2016.07.20
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Abstract

A triode for alternating current (TRIAC) dimming light emitting diode (LED) driver, which adopts a variable switched capacitor for LED dimming and LED power regulation, is proposed in this paper. The proposed LED driver is power efficient, reliable, and long lasting because of the TRIAC switch that serves as its main switch. Similar to previous TRIAC dimmers for lamps, turn-on timing of a TRIAC switch can be controlled by a volume resistor, which modulates the equivalent capacitance of the proposed variable switched capacitor. Thus, LED power regulation against source voltage variation and LED dimming control can be achieved by the proposed LED driver while meeting the global standards for power factor (PF) and total harmonic distortion (THD). The long life and high power efficiency of the proposed LED driver make it appropriate for industrial lighting applications, such as those for streets, factories, parking garages, and emergency stairs. The detailed analysis of the proposed LED driver and its design procedure are presented in this paper. A prototype of 80 W was fabricated and verified by experiments, which showed that the efficiency, PF, and THD at Vs = 220 V are 93.8%, 0.95, and 22.5%, respectively; 65 W of LED dimming control was achieved with the volume resistor, and the LED power variation was well mitigated below 3.75% for 190 V < Vs < 250 V.

Keywords

I. INTRODUCTION

Conventional lamps such as fluorescent lamps and incandescent lamps are being replaced with light emitting diode (LED) lamps due to their high efficacy and long life [1]-[6]. LED drivers provide LED lamps with controlled or regulated current regardless of source voltage or temperature variations. Switched-mode-power-supply (SMPS) LED drivers are most widely used because of their compact size, high power factor (PF), and low total harmonic distortion (THD) [7]-[24]. However, these LED drivers may suffer from switching loss and a relatively short operating life in comparison with LED lamps. These drawbacks reduce the total lifetime of LED lighting systems composed of LED lamps and dedicated LED drivers. Hence, passive LED drivers are preferred because of their extremely high power efficiency and long life [25]-[29]. Even though switching devices such as MOSFET and BJT are not adopted in the LED drivers, the PF and THD characteristics can be satisfied with global standards with a simple structure. However, one of the major drawbacks of passive LED drivers in LED lamps is their lack of a current regulation capability for source voltage variation. This shortcoming hinders the commercialization of such LED drivers.

In order to provide LED power regulation capability with high efficiency and long-life characteristics, a triode for alternating current (TRIAC) dimming control LED driver with a passive input filter was proposed [30], [31]. This LED driver adopts a variable switched capacitor to modulate LED power, which can be controlled through the turn-on duration of a TRIAC together with a diode for alternating current (DIAC); in this way, it was proven that LED dimming and LED power regulation are successfully achieved. Due to the bulky size of two inductors and many capacitors in a passive input filter, however, the total size of this LED driver becomes large, and efficiency is as low as 92%; hence, this LED driver may not be a good candidate for practical lighting solutions. Furthermore, a complicated passive input filter, which is composed of two inductors and five capacitors, makes this LED circuit almost impossible to be analyzed due to its high-order system.

In this paper, a TRIAC dimming LED driver, which reduces the number of passive components in the passive input filter and improves power efficiency in comparison with the aforementioned TRIAC dimming control LED driver [30]-[31], is proposed, as shown in Fig. 1. Only one inductor L1 and two capacitors C1 and C2 with a TRIAC, DIAC, and volume resistor are adopted for the passive input filter and the variable switched capacitor [32], which makes this LED circuit simple, compact, and possible to analyze. The detail operating principle of the proposed variable switched capacitor, design procedure, and additional simulation and experimental results, which were not described in [32] in detail, are additionally provided in this paper. It shows successful LED dimming and LED power regulation capabilities, meeting the PF and THD standards over a wide range of source voltages.

Fig. 1.Overall circuit configuration of the proposed TRIAC dimming LED driver.

 

II. STATIC ANALYSIS OF THE PROPOSED LED DRIVER

A. Operating Principle of the Proposed LED Driver

The proposed LED driver, as shown in Fig. 1, is based on the previous TRIAC dimming LED driver [30], [31]; hence, except for a passive input filter and a variable switched capacitor, the other circuit components such as a DC power supply circuit and a feedback control circuit are the same as previous one. In order to provide this LED driver with an LED power control function, a conventional TRIAC switch with a DIAC is inserted in series with C1 to vary the connecting time portion of C1 in a switching period, which results in a variation of equivalent capacitance for the proposed variable switched capacitor. Note that ns and np are the number of LEDs in series and parallel, respectively, and Cs is used only for the PF compensation.

As shown in Fig. 2, the equivalent circuit of the proposed LED driver can be obtained, by regarding the switched capacitor circuit of C1, C2, and R1 with a TRIAC and DIAC as an equivalent variable capacitor Cv, and by simplifying the diode rectifier and DC load circuit as an equivalent resistor Re [33]-[34]. Note that high-order switching harmonics are neglected, and only the fundamental components of voltage and current are considered. The DC voltage gain GV, which is the ratio of output voltage Vo and source voltage Vs, is then determined as follows:

where α is a DC-AC voltage conversion ratio when a bridge diode is converted to an equivalent auto-transformer [35]-[38].

Fig. 2.The equivalent circuit of the proposed LED driver.

As identified from (1), GV increases when Cv increases as long as the source angular frequency ωs is less than the resonant angular frequency ωr , which is exactly the case in the proposed design, as follows:

Therefore, the LED power, corresponding to Vo, can be appropriately controlled by Cv.

B. Operating Mode Analysis of the Proposed Led Driver

The operating mode of the proposed LED driver can be classified into four modes, as shown in Figs 3-4, where each mode will be described in detail in this section. The LED lamps can be replaced with a dynamic resistance rd and a DC voltage source Vd at a specific operating point of the LED current [29]-[31]. The internal resistance of L1 and the other equivalent series resistances (ESRs) of the capacitors, a TRIAC, a DIAC, and diodes in a diode rectifier are omitted from consideration throughout this paper for simplicity of analysis. The characteristics of all the LED lamps are assumed to be identical, and the temperature distribution over the LED lamps is assumed to be even. All the circuit parameters are assumed to be ideal unless otherwise specified.

Fig. 3.Operating modes of the proposed LED driver.

Fig. 4.Waveform diagrams of the proposed LED driver.

Mode 1 [t0, t1] : As shown in Fig. 4, the phase difference between Vs and Vo ϕs should be defined to identify the beginning of the charging time for C2 t1. The detailed procedure to derive ϕs has been explained for a passive-type LC3 LED driver [29], except for a parallel resonance capacitor, which is not connected to the inductor L1 in parallel in the proposed LED driver.

Mode 2 [t1, t2] : In this mode, the TRIAC is turned off and the initial condition is v2(t1) = 0. After t1, the charging of v2 through L1, C1, and R1 is initiated until v2 reaches the DIAC voltage VD. Given the negative charge of v1 from the previous negative polarity operation of vs, v1 is approximately –VL in this mode. R1 is so largely chosen, i.e., R1 = 1 MΩ, in this paper; hence, it is assumed that DC voltage v1 and AC voltage vs are applied to R1. Then, the current of R1 iR1 can be determined as follows:

Therefore, v2 can be determined as follows:

From (5), the end of the charging time for v2, i.e. t2, can be identified when v2(t2) = VD.

Mode 3 [t2, t3] : In this mode, the TRIAC is turned on and the initial conditions are v1(t2) = vo(t2)= -VL, iL1(t2) = 0, and io(t2) = 0. Thereafter, the proposed LED circuit becomes an LC series resonant circuit, which is composed of L1 and C1, as shown in Fig. 3(c). Then, v1 can be determined as follows:

From (6), iL1 can be determined as follows:

From (6a), the end of this mode can be determined when v1(t3) = VL.

Mode 4 [t3, t4] : In this mode, the TRIAC is turned off. The diode rectifier is conducted, which results in v1(t) = vo(t) = VL; hence, iL1 can be determined as follows:

From (8), iL1 and io in this mode are the same as those in Mode 1, except for the polarity. This mode ends when vs(t4) = 0.

C. Variable Switched Capacitor

The equivalent capacitance of the proposed variable switched capacitor in Fig. 1 can be controlled by modulating the control time Tc, as identified from Figs 3-4. Tc can be modulated by changing the volume resistor R1 and be determined from (5).

To confirm the modulation of Tc by R1, a PSIM simulation is performed for the circuit of Fig. 1 without a feedback control circuit and a DC power supply circuit. The parameter values of Fig. 1 are as follows: L1 = 1.03 H, C1 = 2.0 μF, C2 = 10 nF, VD = 32 V, CL = 100 μF, ns = 85, and np = 4; then, VL = 235 V is determined with the assumption that Vd and rd are 2.7 V and 3.5 Ω, respectively, at the nominal LED current of 80 mA [29]. As shown in Fig. 5, the simulation results are in good agreement with the theoretical analysis of (5), where Tc1, Tc2, and Tc3 are the control times for Vs = 190 V, 220 V, 250 V, respectively. The R1 values greater than 2.9, 3.5, and 4.1 MΩ for Tc1, Tc2, and Tc3, respectively, cannot be adopted in Fig. 5 because v2 in Fig. 4 cannot reach VD as a result of the slow charging time of Tc given a large R1.

It is possible to find out an equivalent variable capacitance Cv with respect to Tc determined by R1, as identified from Fig. 5; hence, the Cv is found by comparing the LED power of the proposed LED driver with that of a LED circuit with only C1 in the proposed variable switched capacitor of Fig. 1. From Fig. 6, it is found that the equivalent capacitance can be appropriately controlled by R1 from the simulation results, where Cv1, Cv2, and Cv3 are the equivalent variable capacitances for Vs = 190 V, 220 V, 250 V, respectively. In Fig. 6, a load resistor RL is connected to the load for a general static analysis of the proposed variable switched capacitor. The value of RL is set to 734 Ω with consideration of VL = 235 V and IL = 0.32 A in (2): an R1 value greater than 1.9, 2.3, and 2.8 MΩ for Cv1, Cv2, and Cv3, respectively, cannot be adopted in this case. From the simulation results of Fig. 6, calculation and simulation results of normalized VL are shown in Fig. 7, where VL1, VL2, and VL3 are the normalized load voltages for Vs = 190 V, 220 V, 250 V, respectively. From (1)-(2), VL (≡Vo/α) can be calculated and α is assumed to be 0.8 for a non-linear diode rectifier [29]. Therefore, it is found from Fig. 7 that the proposed variable switched capacitor can modulate load voltage for LED dimming and power regulation.

Fig. 5.Calculation and simulation results of control time Tc w.r.t. R1.

Fig. 6.Simulation results of the variable switched capacitance Cv w.r.t. R1.

Fig. 7.Calculation and simulation results of normalized VL w.r.t. R1.

 

III. DESIGN OF THE PROPOSED LED DRIVER

As shown in Fig. 1, the proposed LED driver was based on the passive-type LC3 LED driver [29]; hence, circuit parameters for 80 W of power were chosen in a similar way, as listed in Table I.

TABLE ICIRCUIT PARAMETERS OF THE PROPOSED LED DRIVER

With regard to the proposed variable switched capacitor in Fig. 1, a small value of C2 and a large value of R1 are recommended to reduce the size of C2 and the power loss in R1; hence, C2 and R1 are set to 10 nF and 1.0 MΩ, respectively. The worst case for power dissipation in R1 is roughly (2VL)2/R1≅(2‧235)2 /1M≅221mW, which is below 1/4 W. C1 is selected as 2.0 μF, considering the range of the variable switched capacitance. The breakover voltage VD for the DIAC is set to 32 V, considering commercial availability of the DIAC. The component for the TRIAC was a 2N6075AG with a 600 V voltage rating and a 4 A current rating. The load capacitor CL, which reduces the flickering of LED lamps, is set to 100 μF to satisfy the recent international standard, i.e., IEEE 1789-2015 [39]; hence, the maximum percent flicker of the proposed LED driver when R1 = 3.5MΩ and Vs = 250V, as identified from Figs. 4-5, is calculated as 6.38%, which is less than 10% of the international standards. The PF compensation capacitor Cs is 2.68 μF to satisfy the PF regulation [40].

To confirm LED dimming by the volume resistor R1, a PSIM simulation is performed without a feedback control circuit and a DC power supply circuit, as shown in Fig. 8. The internal resistance of inductor, which is 13 Ω for the prototype of the proposed LED driver in Fig. 9, and the other parasitic components are not considered in the simulation verifications. As R1 increases, Tc increases, and the equivalent capacitance of Cv decreases, which results in the decrease in LED power, as identified from Figs. 5-7. In this way, the LED dimming by the volume resistor is achievable for a wide range of source voltages, like a conventional TRIAC dimmer [18]. For a constant LED power PL = 80 W, R1 should be appropriately varied between 1.3 and 3.4 MΩ for a source voltage of 190 V < Vs < 250 V, which is a ± 30 V variation in the rated source voltage of 220 V. Under constant source voltage of Vs = 220 V, LED dimming is achieved with the volume resistor from 120 W to 60 W by the volume resistor.

Fig. 8.Simulation results of LED power w.r.t. R1.

Fig. 9.Prototype of the proposed LED driver.

 

IV. EXPERIMENTAL VERIFICATIONS

As shown in Fig. 9, a prototype of the proposed LED driver for 80 W LED power was fabricated, as listed in Table I. The other circuit parameters of the DC power supply circuit and feedback control circuit, as shown in Fig. 1, can be determined by a design procedure [30], [31]. The proposed LED driver can be used for high LED power applications that require a power level of as high as 80–100 W. Thus, a slightly large inductor L1 is of no practical concern because of the large accommodation space for industrial lighting applications, which usually require highly efficient and long-lasting LED drivers. An inductor L1 in Fig. 9 was fabricated with a silicon steel plate core, and the parasitic resistance of the fabricated inductor L1 was measured as 13.0 Ω. The current rating of the fabricated inductor was 550 mA, and the total weight of the prototype LED driver was measured as 960 g. The inductor L1 and proposed LED driver weighed 930 and 30 g, respectively; hence, the proposed LED driver is lighter and smaller than conventional TRIAC dimming LED drivers weighing 1,390 g [30], [31].

A. LED Dimming

As shown in Fig. 10, the LED dimming of the proposed LED driver without a feedback control circuit [30]-[31] was experimentally verified. The LED power can be controlled by appropriately modulating the volume resistor R1. For example, LED power can be changed from 112 W to 47 W at Vs = 220 V, which matches well with the simulation results in Fig. 9.

Fig. 10.Experimental results of LED power w.r.t. R1.

As shown in Fig. 10, R1 can be set to satisfy PL = 80 W for each source voltage: R1 = 0.83 MΩ, 1.83 MΩ, and 2.6 MΩ for Vs = 190 V, 220 V, and 250 V, respectively. On the basis of these values, the experimental waveforms of vs, vo, v1, vT, v2, iL1, and io were measured, as shown in Figs. 11 and 12, where Tc increased as R1 increased, as anticipated from Fig. 5; Tc was measured as 0.63 ms, 1.21 ms, and 1.90 ms for Vs = 190 V, 220 V, and 250 V, respectively, whose values are in good agreements with the calculation and simulation results in Fig. 5.

Fig. 11.Experimental waveforms of vs, vo, v1, and vT for Vs = 190 V, 220 V, and 250 V at fs = 60 Hz.

Fig. 12.Experimental waveforms of vT, v2, iL1, and io for Vs = 190 V, 220 V, and 250 V at fs = 60 Hz.

B. LED Power Regulation, PF, and THD

The experimental waveforms of v2, vc, iL, and vL were measured, as shown in Fig. 13, where vc is the control voltage of a current mirror in the feedback control circuit [30], [31]. Obviously, vc increased when vs increased to control the charging time of C2; specifically, Vc was measured as -15 V, 8.9 V, and 12.8 V for Vs = 190 V, 220 V, and 250 V, respectively. As a result, load voltage VL and current IL in Fig. 13 were successfully regulated against a wide range of variations in source voltage.

Fig. 13.Experimental waveforms of v2, vc, iL, and vL for Vs = 190 V, 220 V, and 250 V at fs = 60 Hz.

The experimental results of LED power regulation and power efficiency with respect to the source voltage are shown in Fig. 14. The LED power regulation can be implemented by the feedback control circuit [30]-[31]. The LED power variation was mitigated below 3 W for 190 V < Vs < 250 V, with such variation negligible in terms of the change in light brightness. The power efficiency was measured from 93.6% to 94.0% for 190 V < Vs < 250 V, such range is greater than the power efficiency range of conventional LED drivers [13]-[23], [30]-[31]. Over 80% of all the power losses originated from the conduction loss in the fabricated inductor L1; hence, a higher efficiency than that of the previous LED driver was achieved [30]-[31], and such efficiency can be further improved if the internal resistance of L1 is reduced. The other losses included conduction losses of the diode rectifier and volume resistor, BJTs and op-amps in the feedback control circuit, and other ESRs. As shown in Fig. 15, the measured results of PF and THD also satisfy the global standards for 190 V < Vs < 250 V [40]-[41] and are superior to those of conventional LED drivers [13], [20]-[23]. All the measurement results for different source voltages are summarized in Table II.

Fig. 14.Experimental results of PL and η.

Fig. 15.Experimental results of PF and THD.

TABLE IISUMMARY OF EXPERIMENTAL RESULTS FOR SOURCE VOLTAGE VARIATION (190 V < VS < 250 V)

 

V. CONCLUSION

The proposed TRIAC dimming LED driver with a variable switched capacitor was verified in an 80 W LED application. The driver is simple and compact and may thus serve as a practical solution for industrial lighting applications, such as those for streets, factories, parking garages, and emergency stairs. Contrary to SMPS LED drivers using high-frequency switches, the proposed LED driver is equipped with a TRIAC switch that serves as the main switch [7]-[24]. Hence, the proposed driver is more power-efficient, more reliable, and longer lasting than conventional drivers. LED dimming up to 81% is possible by modulating the volume resistor, which value is enough to modulate LED brightness in practical applications. LED power can be successfully regulated within 3.75% for a wide range of 190 V < Vs < 250 V. The measured power efficiency, PF, and THD were 93.8%, 0.95, and 22.5%, respectively, at Vs = 220 V. By virtue of the proposed TRIAC dimming LED driver, a long operating life, high power efficiency, and LED dimming and LED power regulation capabilities can be successfully realized.

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