Journal of Power Electronics

The Korean Institute of Power Electronics (전력전자학회)
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LED Driver with TRIAC Dimming Control by Variable Switched Capacitance for Power Regulation

  • Received : 2014.10.29
  • Accepted : 2014.12.16
  • Published : 2015.03.20
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Abstract

A TRIAC dimming LED driver that can control the brightness of LED arrays for a wide range of source voltage variations is proposed in this paper. Unlike conventional PWM LED drivers, the proposed LED driver adopts a TRIAC switch, which inherently guarantees zero current switching and has been proven to be quite reliable over its long lifetime. Unlike previous TRIAC type LED drivers, the proposed LED driver is composed of an LC input filter and a variable switched capacitance, which is modulated by the TRIAC turn-on timing. Thus, the LED power regulation and dimming control, which are done by a volume resistor in the same way as the conventional TRIAC dimmers, can be simultaneously performed by the TRIAC control circuit. Because the proposed LED driver has high efficiency and a long lifetime with a high power factor (PF) and low total harmonic distortion (THD) characteristics, it is quite adequate for industrial lighting applications such as streets, factories, parking garages, and emergency stairs. A simple step-down capacitive power supply circuit composed of passive components only is also proposed, which is quite useful for providing DC power from an AC source without a bulky and heavy transformer. A prototype 60 W LED driver was implemented by the proposed design procedure and verified by simulation and experimental results, where the efficiency, PF, and THD are 92%, 0.94, and 6.3%, respectively. The LED power variation is well mitigated to below ${\pm}2%$ for 190 V < $V_s$ < 250 V by using the proposed simple control circuit.

Keywords

I. INTRODUCTION

Light emitting diodes (LEDs) are gradually replacing conventional fluorescent lamps and incandescent lights due to their higher efficacy and longer lifespans [1]-[8]. LED drivers provide regulated or controlled currents to LEDs for constant lighting or dimming, regardless of source voltage changes and temperature variations. Switch mode power supply (SMPS)-type LED drivers with pulse width modulation (PWM) switching, in the hundreds of kHz range, are widely used due to their compact filter size and fair total harmonic distortion (THD) characteristics. However, the SMPS-type LED drivers suffer from large switching losses and shorter lifespans than the LED itself due to the high junction temperature of the main switches [9]-[20]. As an alternative solution, passive-type LED drivers are preferred due to their very high efficiency and extremely long lifespan [21]-[24]. However, the inherent drawbacks of the passive-type LED drivers include their lack of current regulation capability and a relatively large filter size, where this filter size problem can be mitigated by virtue of a reduced heat sink and a compact coil design. Therefore, the passive-type LED drivers may not be a good solution for the applications requiring a constant LED current regardless of source voltage variations.

LED dimming is widely used. It is generally achieved with by PWM switches [25]-[28] or TRIAC switches [29]-[32]. LED dimmers composed of a TRIAC and a DIAC with a variable resistor [29]-[32] have many merits such as a simple structure, high reliability, low cost, and high efficiency. However, these conventional TRIAC dimming LED drivers generally have poor THD characteristics because they are directly connected to the primary side of the source line. Though their power factor (PF) can be mitigated by using appropriate power converters [29]-[31], this type of dimmer is not adequate for high power LED applications due to THD and PF problems.

In this paper, a novel TRIAC dimming LED driver with a passive input filter is proposed, as shown in Fig. 1. This LED driver is based on the LC3 passive-type LED driver configuration [24] to achieve a high efficiency and an extremely long lifespan. A dynamically variable switched capacitance is adopted to modulate the TRIAC turn-on timing so that the LED power can be controlled. Unlike the SMPS-type LED drivers, the proposed LED driver uses a TRIAC, which is a proven and quite robust device in home appliances, that inherently guarantees zero current switching (ZCS) with no additional circuits. The LED dimming is done by a volume resistor, which is similar to conventional manual TRIAC dimmers [29]-[32]. The validity of the proposed LED driver design is verified by simulation and experimental results. They show a successful regulation of LED power, and meet PF and THD standards over a wide range of source voltages.

Fig. 1.Overall circuit diagram of the proposed TRIAC dimming LED driver.

 

II. ANALYSIS OF THE PROPOSED TRIAC DIMMING LED DRIVER

A. Operating Principle of the Main Power Circuit in the Steady State

The power circuit of the proposed LED driver, as shown in Fig. 1, is derived from the temperature-robust LC3 LED driver [24], which is a novel passive-type LED driver, that satisfies PF and THD regulations. In order to provide the power control function with the LED driver, a TRIAC switch with a DIAC is inserted in series with C4 to vary the connecting time portion, which results in a variation of the equivalent capacitance, unlike with conventional variable capacitances [33]-[36]. An additional inductor L2 is newly introduced in the proposed LED driver to prevent the TRIAC from having a large inrush current when it is turned on. Note that ns and np are the numbers of LEDs in series and parallel, respectively. In this paper, all of the circuit components are assumed to be ideal unless otherwise specified.

As shown in Fig. 2(a), the simplified equivalent main power circuit of the proposed LED driver can be obtained by regarding the switched capacitor circuit of C4 and the TRIAC as an equivalent variable capacitor Cv and by simplifying the diode rectifier and DC load circuit as an equivalent resistor [23]-[24], [37]-[38] as follows:

where α is the DC to AC voltage conversion ratio when a bridge diode is converted to an equivalent auto-transformer [39]-[42].

Fig. 2.A simplified static circuit of the proposed LED driver, neglecting harmonic components. (a) Simplified equivalent circuit of the proposed LED driver. (b) Further simplified circuit of (a).

In this section, the high-order switching harmonics are neglected from discussions and only the fundamental components of the voltages and currents are considered for simplification.

By applying Thevenin’s theorem to the left part of the circuit in Fig. 2(a) in the steady state at the source angular frequency ωs, the proposed LED circuit can be converted to hat shown in Fig. 2(b), since C1 and C2 are small. Thus, Veand Le are found as follows:

Therefore, the DC voltage gain GV, which is the ratio of the output voltage Vo and the source voltage Vs, is then determined as follows:

As identified from (3), GV increases when Cv increases as long as the source angular frequency ωs is less than the resonant angular frequency ωr, which is exactly the case of the proposed design, i.e.:

The LED dissipating power, which is proportional to Vo, can be accordingly controlled by changing Cv, as shown in Fig. 3.

Fig. 3.The LED power regulation principle by variable switched capacitance.

B. Operating Principle of the TRIAC Dimmer in the Transient State

The method for changing the capacitance Cv by the proposed TRIAC dimmer, as shown in Fig. 1, is explained in this section.

When the TRIAC is completely turned off, as shown in Fig. 4(a), there is no capacitance, assuming the loading effect of R1-C5 is negligible. As a result, Cv = 0. When the TRIAC is completely turned on, as shown in Fig. 4(b), the capacitance becomes Cv = C4. If the TRIAC is appropriately turned on with a delay time of Tc for every half cycle of the power source Ts, as shown in Fig. 5, the equivalent capacitance Cv can be varied as follows:

Fig. 4.Dynamic circuits of the proposed LED driver for a positive half cycle of the vs, assuming the vL is constant due to a large load capacitor. (a) The TRIAC is turned off. (b) The TRIAC is turned on.

Fig. 5.Waveforms of the TRIAC dimmer.

Unfortunately, it is not possible to find an analytical expression of Cv for an arbitrary control time of Tc. This is due to difficulties in finding the dynamics of the fourth-order power circuit. Therefore, the dynamic characteristics and the DC gain of (3) are simulated for each Tc in the subsequent section. Instead, the method for changing the value of Tc is suggested in this section.

TRIAC-Off Mode [t1, t2]: As shown in Fig. 4(a), the DIAC control input voltage v5 is charged, when the TRIAC is turned off, as follows:

In (6), it is assumed that the LED voltage and capacitor voltage are unchanged, i.e. vo = vL = VL and v4 = -VL, respectively. As a result, the TRIAC voltage is determined as vT = vo – v4 ≈ VL – (-VL) = 2VL. Moreover, v4 is nearly constant because R1 is set to be as high as several MΩ. The DIAC triggering voltage VD is set to be 32 V in this paper, which is much less than VL ≈ 210 V. The capacitor voltage v4 has been determined by the previous switching stage, which will be explained in the next few paragraphs.

TRIAC-On Mode [t2, t3]: As v5 reaches VD, the DIAC and TRIAC are turned on, and C5 is immediately discharged so that v5 becomes zero, as shown in Fig. 4(b). At the same time, the capacitor voltage of C4 is resonated with L2 and other filter elements, and its voltage v4 is increased from -VL at t = t2 to VL at t = t3, as shown in Fig. 5. Note that v4 is kept constant until the TRIAC is turned on again at the next negative polarity of source voltage. As soon as v4 reaches VL, the TRIAC is turned off because the time derivative of the capacitor voltage is eventually zero, i.e., Therefore, the TRIAC is kept turned off during [t3, t6], where its voltage becomes negative at t = t5 due to a reversed turning on of the full bridge diode.

The control time Tc, as shown in Fig. 5, can be derived from (6), assuming ic = Ic for a half cycle, as follows:

From (7), Tc can be normalized by R 1C5, as follows:

As shown in Fig. 6, β, which is proportional to Tc, can be controlled by either the manual dimming volume resistor R1 or the control current Ic so that the LED power can eventually be varied.

Fig. 6.An example of theoretical normalized control time Tc /(R1 C5) w.r.t. Ic for various R1 when VD =32 V and VL = 210 V.

C. Operating Principle of the Control Circuit

In order to feed an appropriate Ic of (8) against variations in the LED voltage VL, a control circuit is introduced, as shown in Fig. 7, where the input and output of the proposed circuit are connected to vo and v5 of Fig. 1, respectively.

Fig. 7.Proposed control circuit for LED power regulation.

The proposed control circuit is composed of a negative peak detector for detecting the envelope of the load voltage, a low pass filter for eliminating harmonic voltage ripples higher than the source frequency of 50 Hz or 60 Hz, an integrator to accumulate the control error signals, and a current mirror for converting the control voltage vc to the control current ic.

As shown in Fig. 5, vo and v5 alternatively change every half cycle. Therefore, it is not feasible to derive a transfer function from the proposed control circuit as it is. Instead, the envelope behavior of vo and v5 is considered in this section.

The resistors R6 and R7 constitute a mathematical comparator as follows:

where the constant DC supply voltage Vcc is used to generate the reference voltage Vref. In (9b), a non-linear negative peak detector is approximated to a rectifier for the envelope of each peak voltage .

The low pass filter and integrator have the following transfer functions, respectively:

where a positive offset current is provided through R5 in the low pass filter because the negative peak detector is a current sink.

The current mirror provides the control current Ic of (8), regardless of the polarity, in proportion to vc as follows:

where Vγ and Vσ are the cut-in voltage (≅ 0.6 V) and saturation voltage (≅ 0.2 V) of a bipolar junction transistor, respectively.

As can be seen from (11a) and (11b), the current mirror operates nearly symmetrically w.r.t. v5, which is essential for symmetrical control of each half cycle. In (11e), the DC offset term of (11d) is omitted from the transfer function of Ic (s) of (8) because this offset is relatively small and can be cancelled by the integrator. Therefore, only the change of the mirror current im is considered.

As shown in Fig. 8, the overall closed-loop transfer function of the proposed TRIAC dimming LED driver can be found from (9)-(11) as follows:

Fig. 8.Overall block diagram of the proposed TRIAC dimming LED driver including the control circuit.

In (12), the LED system transfer function Gp (s) has a negative gain as identified from (3) and (8). In other words, Tc increases as Ic increases, which results in decreases of Cv and Vo. The LED voltage VL, which is the final control goal, is indirectly controlled by Vo (≅ αVL) from (12).

In many cases, the power regulation of LED drivers is not necessarily very fast. Thus, the proposed control circuit is designed to be slow enough so that it can be quite robust to switching noise and disturbances. Then, assuming that the low pass filter and LED system are fast enough when compared to the integrator, the loop gain of (12a) becomes as follows:

As identified from (13c), the proposed LED driver is always stable and its first-order time constant can be adjusted by the control gain.

 

III. DESIGN OF THE PROPOSED TRIAC DIMMING LED DRIVER

A. Main Circuit

As shown in Fig. 1, the proposed LED driver includes a passive-type LC3 LED driver [24]; hence, the main power circuit parameters for 60 W of power were chosen in a similar way to those of the references [24], as listed in the Table I.

Table ICIRCUIT PARAMETERS OF THE PROPOSED MAIN POWER CIRCUIT

Concerning the TRIAC dimmer in Fig. 1, a small value of C5 and a large value of R1 are highly recommended for reducing the capacitor size and power loss in R1. Thus, C5 and R1 are chosen as 10 nF and 1.0 ~ 3.0 MΩ, respectively. The worst case power dissipation in R1 is roughly (2VL)2 / R1 ≅ (2·210)2 / 1M ≅ 176.4 mW, which is well below 1/4 W. The value of C4 is selected as 1.0 μF, considering the range of variable switched capacitance. The breakover voltage VD of the DIAC is selected as 32 V, considering the commercial availability of the DIAC.

To confirm LED dimming by a volume resistor R1, a PSIM simulation was performed, as shown in Fig. 9. As R1 increases, Tc increases according to (7), and Cv decreases, which results in a decrease of the LED power, as shown in Fig. 9. In this way, LED dimming by a volume resistor can be achievable for a wide range of source voltages, like a conventional dimming lamp. For a constant LED power PL = 60 W, R1 should be appropriately varied between 1.0 and 3.0 MΩ for a source voltage of 190 V < Vs < 250 V, which is a ± 30 V variation of the rated source voltage of 220 V. For a constant source voltage of Vs = 220 V, the LED power can be varied from 40 W to 80 W.

Fig. 9.Simulation results of the LED power w.r.t. R1 for various Vs.

B. Control Circuit

To sense the envelope of each peak voltage , as shown in Fig. 7, a large value for R6 is recommended for reducing the power loss in the resistor; hence, R6 is set to 4.7 MΩ so that the worst case power dissipation in R6 can be roughly 10 mW. In other words, which is much lower than 1/4 W. In the steady state, the vf2 of (9b) should be zero, assuming that R5 is large enough, so that the input voltage of the integrator can be zero. Thus, for a given VL that is determined from [24], R7 is calculated as 336 kΩ, as follows:

Harmonic voltage components higher than the third harmonic of the source frequency fs are assumed to be eliminated by the low pass filter. In other words, the time constant of the low pass filter, R4C7 is approximately 1 ms, which results in the selection of R4 = 10 kΩ and C7 = 0.1 µF. For Vcc = 15V, R5 is selected as 2.0 MΩ to provide a positive offset current of about 7.5 µA, which is a requisite for maintaining the non-negative polarity of vf3

In the current mirror of Fig. 7, the maximum value for I1 of Fig. 1 is calculated as 0.42 mA from (6) when R1 = 1.0 MΩ and As can be seen from Fig. 9, the maximum value of R1 for 190 V < Vs < 250 V is 2.9 MΩ, which corresponds to almost a third of I1, i.e. 0.14 mA; hence, the maximum control current Ic,max = Im,max becomes 0.28 mA to give this current difference. Then, R2 is approximately calculated from (11d) as 43.2 kΩ, where Vγ = 0.7 V and the maximum value of vc is 13.5 V.

Finally, the integrator is designed, where the time constant of this integrator R3C6 can be determined from (13c). From (9a), K is calculated to be 0.0667, and the system gain Gp0 is experimentally determined to be 2.3 × 105. Therefore, R3C6 becomes 14 ms, and the values of R3 and C6 are selected as 300 kΩ and 47 nF, respectively, considering a sufficiently slow response time of 400 ms.

C. DC Power Supply

To provide the control circuit with supply voltages of Vcc and - Vcc, a new DC power supply circuit is introduced, as shown in Fig. 10. Because the source voltage is very high when compared to the DC supply voltage, a new capacitive voltage divider of C8 and C9 is used. This simple circuit constitutes a step-down capacitive transformer, which may be replaced with a conventional inductive transformer and provides a galvanic isolation from the source voltage. In this paper, it is assumed that Vcc and - Vcc are symmetrical. In other words, C10 = C11, and the zenor diode voltage Vz is the same.

Fig. 10.Proposed capacitive transformer type DC power supply. (a) Original circuit. (b) Thevenin’s equivalent circuit.

By applying Thevenin’s theorem to Fig. 10(a), an equivalent circuit is obtained, as shown in Fig. 10(b), where the parameters are determined as follows:

The analysis of Fig. 10(b) is straightforward if the input current of a half cycle is found for a given Vz. This analysis is already available [43]-[44], assuming that C10 >> Cth and that the full bridge rectifier circuit is replaced with the proposed positive and negative rectifiers. Then, the average current of the positive DC side power supply Ip becomes as follows, neglecting the diode forward voltage drop [44]-[45]:

In (16), Ip0 corresponds to the shorted load circuit where Vz = 0, and Vp0 corresponds to an open load circuit where Ip = 0.

As can be seen from Fig. 11, the average DC current Ip varies as the source voltage changes for Vs = 190 V, 220 V, and 250 V, respectively. When Vs = 190 V, for a given Vz = Vcc = 15 V, Ip becomes its minimum value Iop,min, which should be larger than the required load current of the positive power supply Ip,req. Meanwhile, when Vs = 250 V, Ip becomes its maximum Iop,max, which means that the zenor diode current, which is the difference between Ip and the required load current, becomes large. Because the zenor diode current should be neither zero nor too large, the operating voltage Vop and the operating current Iop are chosen to be half of Vp0 and Ip0 at the normal source voltage of Vs,norm = 220 V, as follows:

Fig. 11.The Vz - Ip curve of the proposed DC power supply circuit.

On the other hand, the minimum DC current Iop,min at the minimum source voltage of Vs,min = 190 V should always be larger than the required load current Ip,req, as follows:

In this paper, the required DC power is identified as Vop = Ccc = 15 V and Ip,req = 1.5 mA at fs = 60 Hz; hence, Vp0 is selected as 30 V. From (17)-(18), C8 and C9 are calculated as 0.099 µF and 0.932 µF, respectively. Hence, C8 and C9 are selected as 0.1 µF and 1.0 µF, considering commercial product availability. To filter out the ripple in the supply voltage Vcc, both C10 and C11 are selected as 10 µF, which is about ten times greater than Cth.

The design parameters of the proposed control circuit and DC power circuit are summarized in Table II.

Table IICIRCUIT PARAMETERS OF THE PROPOSED CONTROL CIRCUIT AND DC POWER SUPPLY

 

IV. EXPERIMENTAL VERIFICATIONS

As shown in Fig. 12, a prototype of the proposed LED driver was fabricated in accordance with the proposed design procedure, as listed in Tables I and II. The core material in the two inductors L1 and L2 of Fig. 12 was silicon steel plate core, and the internal resistance of the fabricated inductors was measured as 7 Ω. Because the proposed LED driver can be used for high LED power applications whose power level is as high as 60 ~ 100 W, two slightly large inductors are of no practical concern due to the large accommodation space of industrial lighting applications. These applications generally require high efficiency and a long lifespan for an LED driver.

Fig. 12.A prototype of the proposed LED driver with two inductors L1 and L2 and an LED array.

A. LED Dimming

As shown in Fig. 13, the LED dimming of the proposed LED driver without a control circuit is experimentally verified. It is well matched with the simulation results of Fig. 9. Slight discrepancies between the experimental and simulation results come mainly from the internal resistances of L1 and L2. The value of PL can be set appropriately by modulating the volume resistor R1. For instance, PL can be changed from 76 W to 40 W at Vs = 220 V, where over 2.7 MΩ of R1 cannot be used for the LED dimming range because v5 in Fig. 5 cannot reach VD within the control time Tc.

Fig. 13.Experimental results of LED power with respect to R1 for various Vs.

As can be seen from Fig. 13, R1 was selected to satisfy PL = 60 W: R1 = 0.75 MΩ, 1.63 MΩ, and 2.4 MΩ for Vs = 190 V, 220 V, and 250 V, respectively. Based on these values, the experimental waveforms of vs, vo, v4, and v5 were measured, as shown in Fig. 14, where Tc increased as Vs increased, as anticipated from Fig. 5.

Fig. 14.The experimental waveforms of vs, vo, v4, and v5 for Vs = 190 V, 220 V, and 250 V at fs = 60 Hz. (a) Vs = 190 V: R1 = 0.75 MΩ. (b) Vs = 220 V: R1 = 1.63 MΩ. (c) Vs = 250 V: R1 = 2.4 MΩ.

B. LED Power Regulation

As shown in Fig. 15, the experimental waveforms of v5, vc, iL, and vL were measured to verify the LED power regulation by the proposed control circuit. When Vs = 190 V, Vc became -15.0 V, which is the minimum value of vc. However, when Vs > 190 V, the LED power regulation began, which is identified by vc > 0. For instance, vc = 8.5 V for Vs = 220 V, and vc = 12.3 V for Vw = 250 V. In this way, PL was found to be well regulated for 190 V < Vs < 250 V by the proposed control circuit. The time constant of the control circuit was measured as 0.4 s, as shown in Figs. 15(e) and (f), which can be varied, as can be seen from (13c).

Fig. 15.The experimental waveforms of v5, vc, iL, and vL for Vs = 190 V, 220 V, and 250 V at fs = 60 Hz. (a) Vs = 190 V: time scale = 5 ms. (b) Vs = 220 V: time scale = 5 ms. (c) Vs = 250 V: time scale = 5 ms. (d) Vs = 190 V: time scale = 1 s. (e) Vs = 220 V: time scale = 1 s. (f) Vs = 250 V: time scale = 1 s.

The LED power and total efficiency of the proposed LED driver were also measured, as shown in Fig. 16(a), where the maximum LED power variation was well mitigated to within ± 2% and the maximum efficiency was between 93.3% and 91.1% for 190 V < Vs < 250 V. This power variation, ± 2%, of the proposed LED driver is quite negligible in practice. As shown in Fig. 16(b), the measured results of the PF and THD satisfy the global standards for 190 V < Vs < 250 V [45]-[46]. Furthermore, all of the harmonics of the source current is satisfy the IEC61000-3-2 class C standard, as shown in Fig. 17. The measurement results for each Vs are summarized in Table III.

Fig. 16.Experimental results of the PL, η, PF, and THD w.r.t. the Vs at fs = 60 Hz. (a) The LED power PL and efficiency η. (b) The PF and THD.

Table IIIMEASURED RESULTS FOR SOURCE VOLTAGE VARIATION (190 V < VS < 250 V)

 

V. CONCLUSION

The proposed TRIAC dimming LED driver by variable switched capacitance has been verified for 60 W LED applications. The LED power can be successfully regulated to within ± 2% for a wide range of 190 V < Vs < 250 V. The measured THD, PF, and power efficiency were 6.3%, 0.94, and 92.3% at Vs = 220 V, respectively. The temperature increase of the LED driver was merely 6℃. With this novel TRIAC dimming LED driver, high efficiency and a long lifespan were achieved, since these are inherent merits of passive-type LED drivers. In addition, LED dimming and LED power regulation were successfully realized. Thus, the proposed LED driver is expected to be utilized in industrial lighting applications such as streets, factories, parking garages, and emergency stairs.

References

  1. M. R. Krames, O. B. Shchekin, R. Mueller-Mach, G. O.Mueller, L. Zhou, G. Harbers, and M. G. Craford, “Status and future of high-power light-emitting-diodes for solid-state lighting,” Journal of Display Technology, Vol. 3, No. 2, pp. 160-175, Jun. 2007. https://doi.org/10.1109/JDT.2007.895339
  2. J. Cardesin, J. Ribas, J. Garcia-Garcia, M. Rico-Secades, A. J. Calleja, E. L. Corominas, and M. A. Dalla Costa, “LED permanent emergency lighting system based on a single magnetic component,” IEEE Trans. Power Electron., Vol. 24, No. 5, pp. 1409-1416, May 2009. https://doi.org/10.1109/TPEL.2009.2013138
  3. D. A. Steigerwald, J. C. Bhat, D. Collins, R. M. Fletcher,M. O. Holcomb, M. J. Ludowise, P. S. Martin, and S. L. Rudaz, “Illumination with solid state lighting technology,” IEEE J. Sel. Topics Quantum Electron., Vol. 8, No. 2, pp. 310–320, Apr. 2002. https://doi.org/10.1109/2944.999186
  4. Matthias Wendt and Jan-Willem Andriesse, “LEDs in real lighting applications: from niche markets to general lighting,” IEEE Industry Applications Conference, pp. 2601-2603, 2006.
  5. S. Y. (Ron) Hui and Y. X. Qin, “A general photo-electro-thermal theory for light emitting diode (LED) systems,” IEEE Trans. Power Electron., Vol. 24, No. 8, pp. 1967-1976, Aug. 2009. https://doi.org/10.1109/TPEL.2009.2018100
  6. Y. K. Cheng and K. W. E. Cheng, “General study for using LED to replace traditional lighting devices,” IEEE International Conference on Power Electronics Systems and Applications (ICPESA), pp. 173-177, 2006.
  7. M. S. Shur and R. Zukauskas, “Solid-state lighting: toward superior illumination,” in Proc. the IEEE, Vol. 93, No. 10, pp. 1691-1703, 2005. https://doi.org/10.1109/JPROC.2005.853537
  8. K. Streubel, N. Linder, R. Wirth, and A. Jaeger, “High brightness AlGaInP light-emitting diodes,” IEEE J. Sel. Topics Quantum Electron., Vol. 8, No. 2, pp. 321-332, Mar. 2002. https://doi.org/10.1109/2944.999187
  9. Q. Hu and R. Zane, “A 0.9 PF LED driver with small LED current ripple based on series-input digitally-controlled converter modules,” IEEE Applied Power Electronics Conference and Exposition (APEC), pp. 2314-2320, 2010.
  10. J. M. Alonso, J. Viña, D. G. Vaquero, G. Martínez, and R. Osorio, “Analysis and design of the integrated double buck-boost converter as a high-power-factor driver for power-LED lamps,” IEEE Trans. Ind. Electron., Vol. 59, No. 4, pp. 1689-1697, Apr. 2012. https://doi.org/10.1109/TIE.2011.2109342
  11. Q. Hu and R. Zane, “LED driver circuit with series-input-connected converter cells operating in continuous conduction mode,” IEEE Trans. Power Electron., Vol. 25, No. 3, pp. 574-582, Mar. 2010. https://doi.org/10.1109/TPEL.2009.2029337
  12. Z. Ye, F. Greenfeld, and Z. Liang, “Single-stage offline SEPIC converter with power factor correction to drive high brightness LEDs,” IEEE Applied Power Electronics Conference and Exposition (APEC), pp. 546-553, 2009.
  13. Y. Hu, L. Huber, and M. M. Jovanovi´, “Single-stage, universal-input AC/DC LED driver with current-controlled variable PFC boost inductor,” IEEE Trans. Power Electron., Vol. 27, No. 3, pp. 1579-1588, Mar. 2012. https://doi.org/10.1109/TPEL.2010.2082564
  14. X. Xie, J. Wang, C. Zhao, Q. Lu, and S. Liu, “A novel output current estimation and regulation circuit for primary side controlled high power factor single-stage flyback LED driver,” IEEE Trans. Power Electron., Vol. 27, No. 11, pp. 4602-4612, Nov. 2012. https://doi.org/10.1109/TPEL.2012.2190523
  15. Y. Zhou, X. Li, X. Ye, and G. Zhai, “A remaining useful life prediction method based on condition monitoring for LED driver,” IEEE conference on Prognostics and System Health Management (PHM), pp. 1-5, 2012.
  16. Q. Hu and R. Zane, “Minimizing required energy storage in off-line LED drivers based on series-input converter modules,” IEEE Trans. Power Electron., Vol. 26, No. 10, pp. 2887-2895, Oct. 2011. https://doi.org/10.1109/TPEL.2010.2087772
  17. Y. Hu and M. M. Jovanovic, “A novel LED driver with adaptive drive voltage,” IEEE Applied Power Electronics Conference and Exposition (APEC), pp. 565-571, 2008.
  18. M. Ali, M. Orabi, M. E. Ahmed, and A. El-Aroudi, “Design consideration of modified SEPIC converter for LED lamp driver,” IEEE Power Electronics for Distributed Generation Systems (PEDG), pp. 394-399, 2010.
  19. D. Gacio, J. M. Alonso, A. J. Calleja, J. Garcia, and M. Rico-Secades, “A universal-input single-stage high-power-factor power supply for HB-LEDs based on integrated buck–flyback converter,” IEEE Trans. Ind. Electron., Vol. 58, No. 2, pp. 589-599, Feb. 2011. https://doi.org/10.1109/TIE.2010.2046578
  20. J. M. Alonso, J. Viña, D. G. Vaquero, G. Martínez, and R. Osorio, “Analysis and design of the integrated double buck–boost converter as a high-power-factor driver for power-LED lamps,” IEEE Trans. Ind. Electron., Vol. 59, No. 4, pp. 1689-1697, Apr. 2012. https://doi.org/10.1109/TIE.2011.2109342
  21. S. Y. R. Hui, S. N. Li, X. H. Tao, W. Chen, and W. M. Ng, “A novel passive offline LED driver with long lifetime,” IEEE Trans. Power Electron., Vol. 25, No. 10, pp. 2665-2672, Oct. 2012. https://doi.org/10.1109/TPEL.2010.2048436
  22. W. Chen, “A comparative study on the circuit topologies for offline passive light-emitting diode (LED) drivers with long lifetime & high efficiency,” IEEE Energy Conversion Congress and Exposition (ECCE), pp. 724-730, 2010.
  23. B. H. Lee, H. J. Kim, and C. T. Rim, “Robust passive LED driver compatible with conventional rapid-start ballast,” IEEE Trans. Power Electron., Vol. 26, No. 12, pp. 3694-3706, Dec. 2011. https://doi.org/10.1109/TPEL.2011.2153875
  24. E. S. Lee, B. H. Choi, J. P. Cheon, B. C. Kim, and C. T. Rim, “Temperature-robust LC3 LED driver with low THD, high efficiency, and long life,” IEEE International Power Electronics Conference (IPEC), pp. 3358-3364, 2014.
  25. Y.-H. Liu, Z.-Z. Yang, and S.-C. Wang, “A novel sequential-color RGB-LED backlight driving system with local dimming control and dynamic bus voltage regulation,” IEEE Trans. Consum. Electron., Vol. 56, No. 4, pp. 2445-2452, Nov. 2010. https://doi.org/10.1109/TCE.2010.5681126
  26. A. Mirvakili and V. Joyner, “A digitally-controlled, bi-level CMOS LED driver circuit combining PWM dimming and data transmission for visible light networks,” IEEE GLOBECOM Workshops (GC Wkshps), pp. 1067-1071, 2010.
  27. W. Feng, F. C. Lee, and P. Mattavelli, “Optimal trajectory control of LLC resonant converters for LED PWM dimming,” IEEE Trans. Power Electron., Vol. 29, No. 2, pp. 979-987, Feb. 2014. https://doi.org/10.1109/TPEL.2013.2257864
  28. D. Gacio, J. M. Alonso, J. Garcia, L. Campa, M. J. Crespo, and M. Rico-Secades, “PWM series dimming for slow-dynamics HPF LED drivers: the high-frequency approach,” IEEE Trans. Ind. Electron., Vol. 59, No. 4, pp. 1717-1727, Apr. 2012. https://doi.org/10.1109/TIE.2011.2130503
  29. J. Zhang, H. Zeng, and T. Jiang, “A primary-side control scheme for high-power-factor LED driver with TRIAC dimming capability,” IEEE Trans. Power Electron., Vol. 27, No. 11, pp. 4619-4629, Nov. 2012. https://doi.org/10.1109/TPEL.2012.2187341
  30. L. Yan, B. Chen, and J. Zheng, “A new TRIAC dimmable LED driver control method achieves high-PF and quality-of-light,” IEEE Applied Power Electronics Conference and Exposition (APEC), pp. 969-974, 2012.
  31. J. T. Hwang, M. S. Jung, D. H. Kim, J. H. Lee, M. H. Jung, and J. H. Shin, “Off-the-line primary side regulation LED lamp driver with single-stage PFC and TRIAC dimming using LED forward voltage and duty variation tracking control,” IEEE J. Solid-State Circuit, Vol. 47, No. 12, pp. 3081-3094, Dec. 2012. https://doi.org/10.1109/JSSC.2012.2225735
  32. R. Zhang and H. S.-H. Chung, “A TRIAC-dimmable LED lamp driver with wide dimming range,” IEEE Trans. Power Electron., Vol. 29, No. 3, pp. 1434-1446, Mar. 2014. https://doi.org/10.1109/TPEL.2013.2263935
  33. J.-H. Cheng, A. F. Witulski, and J. L. Vollin, “A small-signal model utilizing amplitude modulation for the class-D converter at fixed frequency,” IEEE Trans. Power Electron., Vol. 15, No. 6, pp. 1204-1211, Nov. 2000. https://doi.org/10.1109/63.892835
  34. A. Katsuki and Y. Sugimoto, “Design of a distortion-free PFC CV/CC ac power supply having variable capacitance devices,” IEEE Telecommunications Energy Conference (INTELEC), pp. 1-6, 2011.
  35. T. T. Nguyen, M. F. Kandlawala, A. H. Rahim, and M. A. Alam, “Dynamic performance of a grid connected wind generation system with fuzzy logic controlled variable capacitance compensation,” IEEE Australasian Universities Power Engineering Conference (AUPEC), pp. 1-6, 2008.
  36. M. Uenohara, “Noise consideration of the variable capacitance parametric amplifier,” IEEE Proceedings of the IRE, Vol. 48, No. 2, pp. 169-179, 1960. https://doi.org/10.1109/JRPROC.1960.287461
  37. J. Huh, S. W. Lee, W. Y. Lee, G. H. Cho, and C. T. Rim, “Narrow-width inductive power transfer system for online electrical vehicles,” IEEE Trans. Power Electron., Vol. 26, No. 12, pp. 3666-3679, Dec. 2011. https://doi.org/10.1109/TPEL.2011.2160972
  38. S. Lee, B. Choi, and C. T. Rim, “Dynamics characterization of the inductive power transfer system for online electric vehicles by Laplace phasor transform,” IEEE Trans. Power Electron., Vol. 28, No. 12, pp. 5902-5909, Dec. 2013. https://doi.org/10.1109/TPEL.2013.2254500
  39. C. T. Rim and G. H. Cho, “Phasor transformation and its application to the DC/AC analyses of frequency/phase controlled series resonant converters (SRC),” IEEE Trans. Power Electron., Vol. 5, No. 2, pp. 201-211, Apr. 1990. https://doi.org/10.1109/63.53157
  40. C. T. Rim, D. Y. Hu, and G. H. Cho, “Transformers as equivalent circuits for switches: general proofs and D-Q transformation-based analysis,” IEEE Trans. Ind. Appl., Vol. 26, No. 4, pp. 777-785, Jul./Aug. 1990. https://doi.org/10.1109/28.56005
  41. C. T. Rim, “Unified general phasor transformation for AC converters,” IEEE Trans. Power Electron., Vol. 26, No. 9, pp. 2465-2475, Sep. 2011. https://doi.org/10.1109/TPEL.2011.2107920
  42. C. B. Park, S. W. Lee, and C. T. Rim, “Static and dynamic analyses of three-phase rectifier with LC input filter by Laplace phasor transformation,” IEEE Energy Conversion Congress and Exposition (ECCE), pp. 1570-1577, 2012.
  43. N. O. Sokal, K. K. Sum, and D. C. Hamill, “A capacitor-fed, voltage-step-down, single-phase, non-isolated rectifier,” IEEE Applied Power Electronics Conference and Exposition (APEC), pp. 208-215, 1998.
  44. V. M. Rajović and N. S. Jovičić, “The capacitive divider power supply and its design problem,” IEEE 19th Telecommunications Forum (TELFOR), pp. 852-855, 2011.
  45. ENERGY STAR Program Requirements For Solid State Lighting Luminaires, Eligibility Criteria – Version 1.1, 2008.
  46. Limits for Harmonic Current Emissions (Equipment Input Current ≤ 16A per Phase), IEC 61000-3-2 class C Std., 2009.

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