Half-Bridge Zero Voltage Switching Converter with Three Resonant Tanks

Abstract

This paper presents a zero voltage switching (ZVS) converter with three resonant tanks. The main advantages of the proposed converter are its ability to reduce the switching losses on the power semiconductors, decrease the current stress of the passive components at the primary side, and reduce the transformer secondary windings. Three resonant converters with the same power switches are adopted at the low voltage side to reduce the current rating on the transformer windings. Using a series-connection of the transformer secondary windings, the primary side currents of the three resonant circuits are balanced to share the load power. As a result, the size of both the transformer core and the bobbin are reduced. Based on the circuit characteristics of the resonant converter, the power switches are turned on at ZVS. The rectifier diodes can be turned off at zero current switching (ZCS) if the switching frequency is less than the series resonant frequency. Therefore, the reverse recovery losses on the rectifier diodes are overcome. Experiments with a 1.6kW prototype are presented to verify the effectiveness of the proposed converter.

I. INTRODUCTION

High efficiency power converters have drawn a great deal of attention for use in modern power electronic applications such as renewable energy conversion systems, Photovoltaic (PV) inverter systems, industry power conversion systems and personal commercial power units. To meet the efficiency requirements of the Environment Protection Agency (EPA) and the Climate Savers Computing Initiatives (CSCI) for modern power supply units, soft switching converters are generally adopted. Zero voltage switching (ZVS) techniques such as active clamp techniques and full-bridge phase shift pulse-width modulation (PWM), have proposed to reduce the switching losses of MOSFETs. However, the ZVS ranges of these techniques are limited to specific input voltage ranges or load conditions. Series resonant converters and parallel resonant converters have proposed in [1], [2]. They exhibit both high efficiency and low noise. However, the output voltage cannot be properly regulated at the no-load condition in conventional series resonant converters. The LLC series resonant converters in [3]-[11] have been proposed with the advantages of high voltage gain, high conversion efficiency and high power density. All of the power switches can be turned on at ZVS. If the operating switching frequency is lower than the series resonant frequency, the rectifier diodes can be turned off at zero current switching (ZCS). As a result, the reverse recovery losses for the rectifier diodes are reduced.

This paper presents a new resonant converter with three series resonant circuits and a full-wave diode rectifier to achieve soft switching for all of the power switches. Due to the resonant behavior, the power switches can be turned on under ZVS with wide input voltage ranges and load conditions. If the operating switching frequency is lower than the series resonant frequency, the rectifier diodes can be turned off under ZCS. Therefore, the switching losses of the power switches and the reverse recovery problem of the rectifier diodes can be improved. Three resonant circuits with the same power switches are adopted in order to reduce the size of the magnetic cores and the current stress on the primary windings. The secondary windings of two transformers are connected in series in order to balance the primary currents and to share the input current. Finally, the design procedure and test results obtained with a 1.6kW prototype are provided to verify the effectiveness of the proposed converter.

 

II. CIRCUIT CONFIGURATION

A conventional LLC with a half-bridge converter can be adopted in renewable energy DC/DC converters. Basically the output voltage of a renewable energy source is low voltage. As a result, the input voltage of the LLC converter is a low voltage such as 24V-48V, and the output voltage of the LLC is a high voltage such as 400V. In order to reduce the input current stress of the LLC converter, a parallel connection of several LLC converters can be adopted. However, more power switches are needed in this parallel LLC topology. Fig. 1 gives the circuit configuration of the proposed converter with the ZVS/ZCS feature. The proposed converter is controlled by using the frequency modulation technique to regulate the output terminal voltage under input voltage and load current variations. Three resonant tanks with the same power switches are adopted in the proposed converter. The three resonant circuits have the same power switches Q1 and Q2 and secondary side components D1-D4 and Co. The first resonant tank includes Cr1, Lr1 and Lm1. The components of the second resonant tank are Cr2, Lr2 and Lm2. The third resonant tank includes Cr3, Cr4, Lr3 and Lm3. Each resonant tank transfers one-third of the power to the output load. As a result, the current ratings of the resonant inductors, resonant capacitors and transformers are reduced. In order to balance the primary side currents of the three resonant tanks, the secondary windings of T1-T3 are connected in series. Vin and Vo are the input and output side voltages, the power switches Q1 and Q2 make up a half-bridge network, (Cr1, Lr1 and Lm1), (Cr2, Lr2 and Lm2) and (Cr3, Cr4, Lr3 and Lm3) make up the three resonant tanks. T1-T3 are the three isolation transformers. D1-D4 are the rectifier diodes. Coss1 and Coss2 are the output capacitances of Q1 and Q2, respectively. Co is the output capacitance. Based on the resonant behaviors, the power switches Q1 and Q2 are turned on under ZVS within all of the load ranges. If the switching frequency is lower than the series resonant frequency at the full load and maximum input voltage condition, the rectifier diodes D1-D4 can be turned off under ZCS. Therefore, the reverse recovery losses on the rectifier diodes are improved. General fast reverse recovery diodes instead of ultra-fast reverse recovery diodes can be adopted at the output side.

Fig. 1.Circuit configuration of proposed resonant converter.

 

III. OPERATION PRINCIPLE

Before discussing the proposed converter, the following assumptions are made to simplify the system analysis. The transformers T1-T3 are identical with the same magnetizing inductances Lm1=Lm2=Lm3=Lm and turns ratio n=np/ns. The power switches have the same output capacitances Coss1=Coss2=Coss. The four resonant capacitances are Cr1=Cr2=2Cr3=2Cr4=Cr. The three series resonant inductances are identical Lr1=Lr2=Lr3=Lr. A frequency modulation scheme is adopted to regulate output voltage at the desired voltage level. If the switching frequency is greater than the series resonant frequency, there are four operation modes during one switching cycle. However, there are six operation modes in a switching cycle if the switching frequency is less than the series resonant frequency. Since the switching frequency of the proposed converter at full load is designed to be less than the series resonant frequency, there are six operating modes in the proposed converter. The key waveforms of the proposed converter in a switching cycle are given in Fig. 2, and the equivalent circuits of the proposed converter at different operation modes are shown in Fig. 3. Before time t0, Q1 and Q2 are both turned off. The drain voltage vQ1,ds decreases, the drain voltage vQ2,ds increases and D1 and D4 are both conducting.

Fig. 2.Key waveforms of the proposed converter.

Fig. 3.Equivalent circuits of the proposed converter at different operation modes. (a) Mode 1. (b) Mode 2. (c) Mode 3. (d) Mode 4. (e) Mode 5. (f) Mode 6.

Mode 1 [t0≤t

Mode 2 [t1≤t

Mode 3 [t2≤t

Mode 4 [t3≤t

Mode 5 [t4≤t

Mode 6 [t5≤t

At time T+t0, the drain voltage vQ1,ds decreases to zero voltage, and the anti-parallel diode of Q1 is forward biased. Then the operating modes of the proposed converter in one switching cycle are completed.

 

IV. CIRCUIT CHARACTERISTICS

The resonant tank is basically a band-pass filter. The input voltage of the resonant tank is a square wave voltage. If the bandwidth of the band-pass filter is much less than the switching frequency, the harmonics of the input square wave voltage can be neglected at the output of the resonant circuit. A frequency modulation scheme is used to regulate the output voltage. An equivalent circuit of the proposed converter is given in Fig. 4. The duty ratio of each power switch is 0.5 so that the input voltage to the resonant tank is a square waveform. Based on a Fourier series analysis, the input voltages of the resonant tanks are expressed as:

Fig. 4.Equivalent circuit of the proposed converter.

There is a DC voltage level Vin/2 on the input voltages vinput,1 and vinput,2. This DC voltage level is blocked by the resonant capacitors Cr1 and Cr2. Therefore, the input AC voltage components of the three resonant tanks are identical. Since the three resonant tanks have the same circuit components and parameters, only resonant tank 1, as shown in Fig. 5, is discussed to derive the system AC voltage gain. The secondary side current iD is a quasi-sinusoidal current. When the inductor current iLr1>iLm1, rectifier diodes D2 and D3 are conducting and vLm1=-nVo/3. If iLr1

Fig. 5.Equivalent circuit of each resonant tank with fundamental switching frequency.

where θm is the phase angle of the m-th harmonic frequency. The peak fundamental magnetizing inductor voltage is equal to Lm1,f = 4nVo /(3π) . Since the average value of the secondary winding current is equal to the load current, the peak value of the transformer secondary winding current is expressed as:

Therefore, the load resistance Ro reflected to the transformer primary side is given as [2]:

The AC voltage gain of resonant tank 1, shown in Fig. 5, is derived as:

where k=Lr1/Lm1, , , and fs is the switching frequency. The AC voltage gain at the no-load condition (Q=0) and fs=∞ is given in (11).

If the minimum DC voltage gain at the maximum input voltage case is greater than the AC voltage gain at the no-load condition in (11), then the output voltage of the proposed converter can be controlled.

where Vf is the voltage drop across diodes D1-D4. From (12), the minimum turns ratio of transformers T1-T3 is obtained in (13).

 

V. DESIGN EXAMPLE AND EXPERIMENTAL RESULTS

Based on the system analysis in the previous section, a design example of the proposed converter is presented in this section. A laboratory prototype with a 1.6kW rated power was built to verify the effectiveness of the proposed converter. The output voltage and the load current are 400V and 4A. The input voltage range is from 250V to 300V. The selected series resonant frequency fr is 120kHz. The inductance ratio k=Lr/Lm is selected as 0.2.

Step 1: Turns ratio of T1-T3

The minimum DC gain at the maximum input voltage is designed to be unity (at the series resonant frequency). Therefore, the theoretical turns ratio of T1-T3 is given as:

where Vf is the voltage drop on diodes D1-D4. The actual primary and secondary turns used in the prototype circuit are np=33T and ns=30T.

Step 2: DC voltage gain

The actual minimum and maximum DC gains of the proposed converter are:

Step 3: Q value at a full load

Fig. 6 shows the AC voltage gain curves versus the frequency ratio fs/fr with k=0.2. In (16), the maximum DC gain is 1.183. Therefore, the maximum Q at a full load should be less than 0.4, as shown in Fig. 6. Otherwise, no switching frequency exists to regulate the output voltage at the desired voltage level. As a result, the selected Q value at a full load is 0.4 in the prototype.

Fig. 6.AC voltage gain and DC voltage gain at different frequency ratio fs/fr.

Step 4: AC equivalent resistance

Based on (9), the AC equivalent resistance Rac at a full load is given as:

Step 5: Resonant capacitances and inductances

Since and in (10), the resonant capacitances and inductances are derived as:

The selected k=Lr/Lm=1/5. Therefore, the magnetizing inductances of T1-T3 are given as:

Step 6: No load condition

From (13), the minimum turns ratio of the transformers at the no-load condition is obtained as:

The adopted turns ratio of T1-T3 is n=33/30=1.1>nmin. Therefore, the output voltage of the proposed converter can be controlled at the no-load condition.

Step 7: Power semiconductors

The input maximum voltage is 300V and the input DC current is 1600W/300V=5.3A. Therefore, MOSFETs (IRFP460) with a 500V voltage rating and a 20A current rating are used for power switches Q1 and Q2. The output voltage is 400V and the load current is 4A. FSF10A60 with a 600V voltage rating and a 10A current rating are used for rectifier diodes D1-D4 in the prototype circuit.

Experiments based on a laboratory prototype with the circuit parameters derived in the previous section are provided to verify the effectiveness of the proposed converter. Fig. 7 shows the measured waveforms of the gate voltage, drain voltage and switch current of Q1 and Q2. Before switches Q1 and Q2 are turned on, the drain voltage is decreased to zero. Therefore, switches Q1 and Q2 are turned on at ZVS. Fig. 8 shows the measured waveforms of the gate voltage vQ1,gs and the inductor currents iLr1-iLr3 at the 25% load and full load conditions. It is clear that the three resonant inductor currents iLr1-iLr3 are well balanced. Fig. 9 shows the measured resonant capacitor voltages vCr1-vCr4 at the 25% load and full load conditions. Fig. 10 shows the measured diode currents at the full load condition. When switch Q1 is on, the transformer secondary winding current is is positive and diodes D1 and D4 are conducting. If switch Q1 is off, the transformer secondary winding current is is negative and diodes D2 and D3 are conducting. It is clear that D1-D4 are turned off under ZCS. The curves of the switching frequency and efficiency at different loads are shown in Fig. 11. The measured switching frequencies of the proposed converter at 25%, 50% and 100% loads with a 300V input voltage are 132kHz, 124kHz and 118kHz, respectively. The measured circuit efficiencies of the proposed converter at 25%, 50% and 100% loads with a 300V input voltage are 89.7%Hz, 93.3% and 92.1%, respectively.

Fig. 7.Measured gate voltage, drain voltage and switch current of Q1 and Q2 at (a) 25% (b) 100% full load conditions.

Fig. 8.Measured gate voltage vQ1,gs, and inductor currents iLr1-iLr3 at (a) 25% load (b) full load.

Fig. 9.Measured resonant capacitor voltages vCr1-vCr4 at (a) 25% load (b) full load.

Fig. 10.Measured waveforms of diode currents at full load condition.

Fig. 11.Measured switching frequency and efficiency of the proposed converter at different loads. (a) Switching frequency versus load. (b) Efficiency versus load.

 

VI. CONCLUSIONS

A new ZVS converter with three resonant circuits with the same power switches is proposed to achieve the functions of ZVS turn-on of the power switches, ZCS turn-off of the rectifier diodes and less current stress for the primary inductors. As a result, the switching losses on the power switches are reduced, and the reverse recovery current on the rectifier diodes is overcome. Three resonant circuits with the same power switches are adopted at the primary side, and each resonant circuit delivers one-third of the load power to the output side. The secondary windings of the three transformers are connected in series to balance the three primary side currents. A full-wave diode rectifier is adopted at the high voltage side to limit the voltage stress of the rectifier diode at the output voltage. A frequency modulation scheme is adopted to derive the AC voltage conversion ratio. Finally, experimental results are presented to verify the effectiveness of the converter.